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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual

Buck Controller with 17μA Quiescent Current

General Description

Beneﬁts and Features

● Meets Stringent Automotive OEM Module Power-

Consumption and Performance Specifications

• 17µA Quiescent Current in Skip Mode

The MAX20034 is an automotive 2.2MHz, dual synchro-

nous step-down controller IC that provides two high-

voltage, synchronous step-down controllers that operate

180° out-of-phase. The IC operates with a 3.5V to 42V

input-voltage supply and can function in dropout condition

by running at 99% duty cycle. It is intended for applications

with mid- to high-power requirements that perform at a

wide input voltage range, such as during automotive cold-

crank or engine stop-start conditions.

• ±1.5% Output-Voltage Accuracy: 5.0V/3.3V Fixed,

or Adjustable Between 1V and 10V

● Enables Crank-Ready Designs

• Wide 3.5V to 36V Input Supply Range

● EMI Reduction Features Reduce Interference with

Sensitive Radio Bands without Sacrificing Wide Input

Voltage Range

The IC’s step-down controllers operate at up to 2.2MHz

frequency to allow small external components, reduced

output ripple, and to guarantee no AM band interference.

The switching frequency is resistor adjustable (220kHz

to 2.2MHz). SYNC input programmability enables three

frequency modes for optimized performance: forced

fixed-frequency operation, skip mode with ultra-low

quiescent current, and synchronization to an external

clock. The IC is also available with spread-spectrum

frequency modulation to minimize EMI interference.

• 50ns (typ) Minimum On-Time Guarantees Skip-

Free Operation for 3.3V Output from a Car Battery

at 2.2MHz

• Spread-Spectrum Option

• Frequency-Synchronization Input

• Resistor-Programmable Frequency Between

220kHz and 2.2MHz

● Integration and Thermally Enhanced Packages Save

Board Space and Cost

The IC features

a

power-OK monitor, overvolt-

• Dual, Up to 2.2MHz Step-Down Controllers

• 180° Out-of-Phase Operation

• Current-Mode Controllers with Forced-PWM

(FPWM) and Skip Modes

age lockout, and undervoltage lockout. Protection

features include cycle-by-cycle current limit and

thermal shutdown. The MAX20034 is specified

for operation over the -40°C to +125°C automotive

temperature range.

• Thermally Enhanced, 28-Pin TQFN-EP Package

● Protection Features Improve System Reliability

• Supply Overvoltage and Undervoltage Lockout

• Overtemperature and Short-Circuit Protection

Applications

● POL Applications for Automotive Power

● Distributed DC Power Systems

● Navigation and Radio Head Units

Ordering Information appears at end of data sheet.

19-100159; Rev 3; 8/19

MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Simpliﬁed Block Diagram

PGOOD1 COMP1

DC-DC1

CONTROL LOGIC

PGOOD LOW LEVEL

PGOOD HIGH LEVEL

PGOOD

COMP

MAX20034

FB1

FEEDBACK-

SELECT LOGIC

EAMP1

INTERNAL

SOFT-START

EN1

BST1

DH1

LX1

V

REF

= 1V

PWM1

CLK1

ZX1

OUT1

CS1

PWM1

80mV (TYP) MAX

DIFFERENTIAL INPUT

STEP-DOWN DC-DC1

CSA1

GATE-DRIVE

LOGIC

DL1

CL

ZERO-

CROSS

COMP

SLOPE-

COMP LOGIC

CURRENT-LIMIT

THRESHOLD

EN1

PGND1

LX1

CLK1

FOSC

OSCILLATOR

IN

SPREAD-SPECTRUM

OPTION AVAILABLE WITH

INTERNAL CLOCK ONLY

EXTERNAL

CLOCK INPUT

BIAS

INTERNAL LINEAR

REGULATOR

CONNECTED HIGH (PWM MODE)

CONNECTED LOW (SKIP MODE)

FSYNC

AGND

FSYNC-SELECT LOGIC

IF 3.25V <

< 5.2V

V

EXTVCC

SWITCHOVER

CLK 180°

OUT-OF-PHASE

CLK2

EN2

PWM2

CLK2

ZX2

BST2

DH2

COMP2

FB2

OUT2

CS2

STEP-DOWN DC-DC2

LX2

DC-DC2 CONTROL LOGIC

SAME AS DC-DC1 ABOVE

GATE-DRIVE

LOGIC

EN2

DL2

PGOOD2

PGND2

LX2

EP

Maxim Integrated

│ 2

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Absolute Maximum Ratings

IN, EN1, EN2, LX_ to PGND.................................-0.3V to +42V

OUT1, OUT2 to AGND..........................................-0.3V to +12V

CS1 to OUT1........................................................-0.3V to +0.3V

CS2 to OUT2........................................................-0.3V to +0.3V

BIAS, FSYNC, PGOOD_, FB_,

DH_ to LX_ (Note 1)................................ -0.3V to V

PGND_ to AGND..................................................-0.3V to +0.3V

+ 0.3V

BST_

Continous Power Dissipation (T = +70°C)

A

28 TQFN (derate 37mW/°C above +70°C)................2285mW

Operating Temperature Range......................... -40°C to +125°C

Junction Temperature......................................................+150°C

Storage Temperature Range............................ -65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

Soldering Temperature (reflow).......................................+260°C

EXTVCC to AGND...............................................-0.3V to +6V

COMP_, FOSC to PGND_.......................-0.3V to V

DL_ to PGND_ (Note 1)...........................-0.3V to V

+ 0.3V

BIAS

BST_ to LX_ (Note 1)..............................................-0.3V to +6V

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these

or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect

device reliability.

Note 1: Self-protected against transient voltages exceeding these limits for ≤ 50ns under normal operation and loads up to the

maximum rated output current.

Package Information

PACKAGE TYPE: 28-PIN TQFN

Package Code

T2855Y-5C

21-100130

90-0027

Outline Number

Land Pattern Number

THERMAL RESISTANCE, FOUR-LAYER BOARD:

Junction to Ambient (θ

)

35°C/W

3°C/W

JA

Junction to Case (θ

)

JC

For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,

“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing

pertains to the package regardless of RoHS status.

Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board.

For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.

Electrical Characteristics

(V

= 14V, C

= 6.8μF, R

= 12kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at

IN

BIAS

FOSC A J

T

= +25°C. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range and relevant supply

A

A A

voltage range are guaranteed by design and characterization.)

PARAMETER

SYMBOL

CONDITIONS

Normal operation

t < 1s

MIN

TYP

MAX

36

UNITS

3.5

Supply Voltage Range

V

IN

42

V

= V

= 0V

6.5

25

10

EN1

EN2

V

= 5V, V

= 0V,

EN1

OUT1

= 5V (no switching)

EN2

40

28

Supply Current

I

µA

IN

EXTVCC

V

= 5V, V

= 3.3V, V

= 0V,

EN2

OUT2

= 3.3V (no switching)

EN1

17

5

EXTVCC

Buck 1 Fixed-Output

Voltage

V

= V , V = 5V, PWM mode

BIAS OUT1

4.925

3.25

5.075

3.35

V

OUT1

FB1

FB2

Buck 2 Fixed-Output

Voltage

V

= V

, V

= 3.3V, PWM mode

3.3

OUT2

BIAS OUT2

Maxim Integrated

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Electrical Characteristics (continued)

(V

= 14V, C

= 6.8μF, R

= 12kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at

IN

BIAS

FOSC A J

T

= +25°C. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range and relevant supply

A

A A

voltage range are guaranteed by design and characterization.)

PARAMETER

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

Output Voltage-Adjustable

Range

Buck 1, Buck 2

PWM

1

10

V

Regulated Feedback

Voltage

V

I

V

0.995

1.005

0.01

470

1.015

1

V

FB1, FB2

Feedback Leakage

Current

I

T

= +25°C

µA

FB1, FB2

A

Feedback Line-

Regulation Error

V

= 3.5V to 36V, V

= 1V

%/V

µS

IN

FB_

Transconductance (from

FB1, 2 to COMP1, 2)

g

= 1V, V = 5V

BIAS

300

97

700

m

FB_

DL_ low to DH_ high

DH_ low to DL_ high

Buck 1, Buck 2

15

99

50

Dead Time

ns

Maximum Duty Cycle

Minimum On-Time

%

t

Buck 1, Buck 2

ns

ON,MIN

PWM Switching-

Frequency Range

f

Programmable

0.22

2

2.2

2.4

MHz

SW

Switching-Frequency

Accuracy

R

= 12kΩ, V

= 5V, 3.3V

2.2

FOSC

BIAS

CS_ Current-Limit

Voltage Threshold

V

LIMIT1,

LIMIT2

V

- V

; V

= 5V, V ≥ 2.5V

OUT

68

3

80

5

92

8

mV

ms

deg

µA

Ω

CS_

OUT BIAS

Soft-Start Ramp Time

Buck 1 and Buck 2

Phase Shift Between

Buck 1 and Buck 2

PWM operation (Note 2)

180

0.001

3

LX1, LX2 Leakage Current

V

= V

or V , T = +25°C

1

6

LX_

PGND_

IN

A

DH1, DH2 Pullup

Resistance

V

= 5V, l

= -100mA

= 100mA

BIAS

DH

DH1, DH2 Pulldown

Resistance

V

= 5V, l

1.5

3

6

3

Ω

BIAS

DH

DL1,2 Pullup Resistance

= 5V, l = -100mA

DL

DL1, DL2 Pulldown

Resistance

= 5V, l = 100mA

1.5

DL

Output Overvoltage

Threshold

Detected with respect to V

rising

105

108

3

112

%

FB_

Output Overvoltage

Hysteresis

Maxim Integrated

│ 4

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Electrical Characteristics (continued)

(V

= 14V, C

= 6.8μF, R

= 12kΩ, T = T = -40°C to +125°C, unless otherwise noted. Typical values are at

IN

BIAS

FOSC A J

T

= +25°C. Limits are 100% tested at T = +25°C and T = +125°C. Limits over the operating temperature range and relevant supply

A

A A

voltage range are guaranteed by design and characterization.)

PARAMETER

SYMBOL

PGOOD_R

PGOOD_F

CONDITIONS

MIN

92

TYP

94

MAX

97

UNITS

Percentage of V

, rising

OUT_

PGOOD Threshold

%

, falling

90

92

95

Leakage Current

Output Low Voltage

Debounce Time

FSYNC INPUT

V

= 5V, T = +25°C

0.01

1

µA

V

PGOOD_

A

I

= 1mA

0.2

SINK

Fault detection, rising and falling

20

µs

Minimum sync pulse > (1/FSYNC - 1/FOSC)

1.8

2.6

MHz

kHz

R

= 12kΩ

FOSC

FSYNC Frequency Range

Minimum sync pulse > (1/FSYNC - 1/FOSC)

= 70kΩ

250

1.4

550

R

FOSC

High threshold

Low threshold

FSYNC Switching

Thresholds

V

0.4

INTERNAL LDO BIAS

Internal BIAS Voltage

V

> 6V, no load

5

V

IN

rising

falling

3.1

2.6

3.25

BIAS

BIAS UVLO Threshold

2.35

3.25

2.85

EXTVCC Operating

Range

5.5

V

EXTVCC Threshold

V

EXTVCC rising, hysteresis = 110mV

3

3.25

TH,EXTVCC

THERMAL OVERLOAD

Thermal-Shutdown

Temperature

(Note 2)

170

20

°C

Thermal-Shutdown

Hysteresis

ENABLE LOGIC INPUT

High Threshold

EN1, EN2

1.8

V

Low Threshold

0.8

1

EN_ Input Bias Current

EN1, EN2 logic inputs only, T = +25°C

0.01

µA

A

Note 2: Guaranteed by design, not production tested.

Maxim Integrated

│ 5

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Typical Operating Characteristics

(T = +25°C, unless otherwise noted.)

A

SHUTDOWN SUPPLY CURRENT

vs. SUPPLY VOLTAGE

STARTUP INTO LOAD

STARTUP INTO NO-LOAD

toc02

toc01

toc03

15

12

9

5V/div

2V/div

V

V_{EN_}

EN_

EN1 = EN2

FSYNC = 0

2V/div

5V/div

V_EXTVCC= V_OUT1

I_OUT1= I_OUT2= 0A

V

V_OUT1

OUT1

5V/div

EN1 = EN2

FSYNC = 0

V

V_PGOOD1

PGOOD1

I

V

= I

= 3A

OUT1

OUT1 OUT2

= V

EXTVCC

6

2V/div

5V/div

2V/div

5V/div

V

V_OUT2

OUT2

3

V

V_PGOOD2

PGOOD2

2ms/div

0

6

9

12 15 18 21 24 27 30 33 36

SUPPLY VOLTAGE (V)

QUIESCENT CURRENT

vs. SUPPLY VOLTAGE

BUCK1 EFFICIENCY vs OUTPUT CURRENT

(f_SW= 2.2MHz)

toc04

toc05

70

60

50

40

30

20

10

0

100

90

80

70

60

50

40

30

20

10

0

ONLY BUCK 1 ON, V_EXTVCC= V_OUT1

ONLY BUCK 2 ON, V_EXTVCC= V_OUT2

BOTH BUCKS ON, V_EXTVCC= V_OUT1

SKIP ENABLED

FPWM

EN1 = HIGH

EN2 = LOW

EXTVCC = V_OUT1

V_OUT1= 5V

0.001

0.01

0.1

1

10

OUTPUT CURRENT (A)

6

12

18

24

30

36

SUPPLY VOLTAGE (V)

BUCK 2 EFFICIENCY vs. OUTPUT CURRENT

(f_SW= 400kHz)

BUCK 1 EFFICIENCY vs. OUTPUT CURRENT

(f_SW= 400kHz)

toc07

toc06

100

90

80

70

60

50

40

30

20

10

0

100

90

80

70

60

50

40

30

20

10

0

SKIP ENABLED

FPWM

SKIP ENABLED

FPWM

EN1 = HIGH

EN2 = HIGH

V_EXTVCC= V_OUT1

V_OUT2= 3.3V

EN1 = HIGH

EN2 = LOW

V_EXTVCC= V_OUT1

V_OUT1= 5V

0.001

0.01

0.1

1

10

0.001

0.01

0.1

1

10

OUTPUT CURRENT (A)

Maxim Integrated

│ 6

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Typical Operating Characteristics (continued)

(T = +25°C, unless otherwise noted.)

A

SWITCHING FREQUENCY

vs. LOAD CURRENT

SWITCHING FREQUENCY

vs. AMBIENT TEMPERATURE

SWITCHING FREQUENCY vs. R_FOSC

toc08

toc10

2.22

toc09

2.22

2.21

2.20

2.19

2.18

2.17

2.16

2.15

2.14

2.13

2.12

2.6

2.4

2.2

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.21

2.20

2.19

2.18

2.17

2.16

2.15

BUCK1

BUCK2

FSYNC = 1

2.14

2.13

2.12

0

1

2

3

4

5

LOAD CURRENT (A)

0

20

40

60

80 100 120 140 160

-40 -25 -10

5

20 35 50 65 80 95 110 125

AMBIENT TEMPERATURE (°C)

R_FOSC(kΩ)

FSYNC SYNCHRONIZATION

OUT-OF-PHASE OPERATION

toc11

toc12

5V/div

V_FSYNC

V

FSYNC

1.8MHz, 50% DUTY-CYCLE SIGNAL ON FSYNC

I_OUT1= I_OUT2= 2.5A

2.2MHz, 50% DUTY-CYCLE SIGNAL ON FSYNC

= I = 2.5A

I

OUT1 OUT2

V_LX1

10V/div

V

10V/div

LX1

V_LX2

10V/div

V

10V/div

LX2

1µs/div

LOAD TRANSIENT RESPONSE (BUCK 2)

LOAD-TRANSIENT RESPONSE (BUCK 1)

toc14

toc13

I_OUT1

5A

5A/div

5A

0A

I_OUT2

5A/div

0A

I_LX

5A/div

10V/div

V_{LX_}

V_LX

V_OUT2= 3.3V

FPWM

EXTVCC = V_OUT2

V_OUT

V_{OUT_}

200mV/

div

200mV/div

V_OUT1= 5V

FPWM

V_EXTVCC= V_OUT1

200µs/div

Maxim Integrated

│ 7

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Typical Operating Characteristics (continued)

(T = +25°C, unless otherwise noted.)

A

OUTPUT LOAD REGULATION

(BUCK 2)

OUTPUT LOAD REGULATION

(BUCK 1)

OUTPUT LINE REGULATION (BUCK 1)

toc15

toc16

toc17

5.10

5.02

5.01

5.00

4.99

4.98

4.97

4.96

3.366

3.333

3.300

3.267

3.234

SKIP

FPWM

I_OUT= 0A

I_OUT= 0.1A

I_OUT= 0.5A

SKIP

FPWM

V_EXTVCC= V_OUT1

5.05

V_EXTVCC= V_OUT1

5.00

4.95

4.90

I_OUT= 1A

I_OUT= 2A

I_OUT= 3A

I_OUT= 4A

FPWM

V_EXTVCC= V_OUT1

0.001

0.01

0.1

1

10

6

9

12 15 18 21 24 27 30 33 36

0.001

0.01

0.1

1

10

INPUT VOLTAGE (V)

OUTPUT CURRENT (A)

COLD CRANK (BUCK 1)

OUTPUT LINE REGULATION (BUCK 2)

toc19

toc18

3.300

3.295

3.290

3.285

3.280

3V/div

1V/div

V_IN

V_OUT

I_OUT= 0A

I_OUT= 0.1A

I_OUT= 0.5A

I_OUT= 1A

I_OUT= 2A

I_OUT= 3A

I_OUT= 4A

I_LX

2A/div

5V/div

V_IN= 14V TO 3.5V TO 5V TO 14V

FPWM, I_OUT= 4A, V_EXTVCC= V_OUT1

FPWM

V_EXTVCC= V_OUT2

V_PGOOD1

6

9

12 15 18 21 24 27 30 33 36

INPUT VOLTAGE (V)

100ms/div

COLD CRANK (BUCK 2)

LOAD DUMP (BUCK 1)

toc20

toc21

3V/div

10V/div

1V/div

V_IN

V_OUT

1V/div

I_LX

2A/div

5V/div

2A/div

5V/div

V_IN= 14V TO 3.5V TO 5V TO 14V

FPWM, I_OUT= 4A, V_EXTVCC= V_OUT2

FPWM, I_OUT= 4A, V_EXTVCC= V_OUT1

V_PGOOD

100ms/div

50ms/div

Maxim Integrated

│ 8

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Typical Operating Characteristics (continued)

(T = +25°C, unless otherwise noted.)

A

SLOW INPUT-VOLTAGE RISING (BUCK 2)

LOAD DUMP (BUCK 2)

SLOW INPUT-VOLTAGE RISING (BUCK 1)

toc24

toc22

toc23

V

10V/div

1V/div

10V/div

1V/div

V_IN

IN

V_OUT

V

= 0 TO 14V

IN

1V/div

FPWM, I

V

= 4A

OUT

= V

OUT1

EXTVCC

I_LX

2A/div

5V/div

V

V_IN= 0 TO 14V

FPWM, I_OUT= 4A

V_EXTVCC= V_OUT1

OUT

2A/div

5V/div

2A/div

5V/div

FPWM, I_OUT= 4A, V_EXTVCC= V_OUT1

I

LX

V_PGOOD

V

PGOOD

5s/div

50ms/div

5s/div

LINE TRANSIENT OUT OF DROPOUT

(BUCK 1)

LINE TRANSIENT OUT OF DROPOUT

(BUCK 2)

toc25

toc26

3V/div

3.42V

V_IN= 5V TO 14V

FPWM, NO LOAD

5.13V

300mV/div

100mV/div

V_LX

5V/div

1A/div

5V/div

1A/div

V_IN= 5V TO 14V

50µs/div

FPWM, NO LOAD

I_LX

50µs/div

SHORT CIRCUIT AFTER REGULATION

(BUCK 1)

SHORT CIRCUIT AFTER REGULATION

(BUCK 2)

toc27

toc28

2V/div

V_OUT1

2V/div

2A/div

5V/div

2A/div

5V/div

V_PGOOD

I_LX

10V/div

50µs/div

V_LX

10V/div

400µs/div

Maxim Integrated

│ 9

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Pin Conﬁguration

TOP VIEW

21 20 19 18 17 16 15

14

13

LX2 22

DH2 23

FSYNC

PGOOD2

12 PGOOD1

24

25

26

27

28

BST2

EN2

MAX20034

IN

11

10

9

EN1

EXTVCC

AGND

BIAS

BST1

DH1

EP

6

+

8

1

2

3

4

5

7

TQFN

(5mm x 5mm)

Pin Description

NAME

FUNCTION

Inductor Connection for Buck 1. Connect LX1 to the switched side of the inductor. LX1 serves as

the lower supply rail for the DH1 high-side gate driver.

1

LX1

2

3

DL1

Low-Side Gate-Driver Output for Buck 1. DL1 output voltage swings from V

Power Ground for Buck 1

to V

.

PGND1

BIAS

PGND1

Positive Current-Sense Input for Buck 1. Connect CS1 to the positive terminal of the current-sense

element. See the Current Limiting and Current-Sense Inputs (OUT_ and CS_) and Current-Sense

Measurement sections.

4

CS1

Output Sense and Negative Current-Sense Input for Buck 1. When using the internal preset 5V

feedback-divider (FB1 = BIAS), the controller uses OUT1 to sense the output voltage. Connect

OUT1 to the negative terminal of the current-sense element. See the Current Limiting and Current-

Sense Inputs (OUT_ and CS_) and Current-Sense Measurement sections.

5

OUT1

Feedback Input for Buck 1. Connect FB1 to BIAS for the 5V ﬁxed output or to a resistive divider

between OUT1 and AGND to adjust the output voltage between 1V and 10V. In adjustable version,

FB1 regulates to 1V (typ). See the Setting the Output Voltage in Buck Converters section.

6

7

8

FB1

COMP1

BIAS

Buck 1 Error-Ampliﬁer Output. Connect an RC network to COMP1 to compensate.

5V Internal Linear Regulator Output. Bypass BIAS to PGND with a low-ESR ceramic capacitor of

6.8µF minimum value. BIAS provides the power to the internal circuitry and external loads. See the

Fixed 5V Linear Regulator (BIAS) section.

9

AGND

Signal Ground for IC

Switchover Comparator Input. Connect a voltage between 3.25V and 5.5V to EXTVCC to power the

IC and bypass the internal bias LDO.

10

EXTVCC

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Pin Description (continued)

NAME

FUNCTION

11

IN

Supply Input. Bypass IN with enough capacitors to supply the two out-of-phase buck converters.

Open-Drain Power-Good Output for Buck 1. PGOOD1 is low if OUT1 is more than 92% (typ)

below the normal regulation point. PGOOD1 asserts low during soft-start and in shutdown.

PGOOD1 becomes high impedance when OUT1 is in regulation. To obtain a logic signal, pull

PGOOD1 up with an external resistor connected to a positive voltage lower than 5.5V.

12

13

14

PGOOD1

PGOOD2

FSYNC

Open-Drain Power-Good Output for Buck 2. PGOOD2 is low if OUT2 is more than 92% (typ)

below the normal regulation point. PGOOD2 asserts low during soft-start and in shutdown.

PGOOD2 becomes high impedance when OUT2 is in regulation. To obtain a logic signal, pull

PGOOD2 up with an external resistor connected to a positive voltage lower than 5.5V.

External Clock-Synchronization Input. Synchronization to the controller operating-frequency ratio

is 1. See the Switching Frequency/External Synchronization section. For FSYNC high, and T

<

ON

T

, ensure there is at least 50μA (including the resistor-divider current on V

) of load

OUT1,2

ON,MIN

current if V

- V

> 1.3V.

BIAS

OUT

Frequency-Setting Input. Connect a resistor from FOSC to AGND to set the switching frequency of

the DC-DC converters.

15

16

FOSC

COMP2

Buck 2 Error-Ampliﬁer Output. Connect an RC network to COMP2 to compensate buck converter 2.

Feedback Input for Buck 2. Connect FB2 to BIAS for the 3.3V ﬁxed output or to a resistive divider

between OUT2 and AGND to adjust the output voltage between 1V and 10V. In adjustable version,

FB2 regulates to 1V (typ). See the Setting the Output Voltage in Buck Converters section.

17

18

19

FB2

OUT2

CS2

Output Sense and Negative Current-Sense Input for Buck 2. When using the internal preset 3.3V

feedback-divider (FB2 = BIAS), the buck uses OUT2 to sense the output voltage. Connect OUT2

to the negative terminal of the current-sense element. See the Current Limiting and Current-Sense

Inputs (OUT_ and CS_) and Current-Sense Measurement sections.

Positive Current-Sense Input for Buck 2. Connect CS2 to the positive terminal of the current-sense

element. See Current Limiting and Current-Sense Inputs (OUT_ and CS_) and Current-Sense

Measurement sections.

20

21

PGND2

DL2

Power Ground for Buck 2

Low-Side Gate-Driver Output for Buck 2. DL2 output voltage swings from V

to V

.

PGND2

BIAS

Inductor Connection for Buck 2. Connect LX2 to the switched side of the inductor. LX2 serves as

the lower supply rail for the DH2 high-side gate driver.

22

23

LX2

DH2

High-Side Gate-Driver Output for Buck 2. DH2 output voltage swings from V

to V

.

LX2

BST2

Boost Flying-Capacitor Connection for High-Side Gate Voltage of Buck 2. Connect a high-voltage

diode between BIAS and BST2. Connect a ceramic capacitor between BST2 and LX2. See the

High-Side Gate-Driver Supply (BST_) section.

24

BST2

25

26

EN2

EN1

High-Voltage-Tolerant, Active-High Digital Enable Input for Buck 2. Driving EN2 high enables Buck 2.

High-Voltage-Tolerant, Active-High Digital Enable Input for Buck 1. Driving EN1 high enables Buck 1.

Boost Flying-Capacitor Connection for High-Side Gate Voltage of Buck 1. Connect a high-voltage

diode between BIAS and BST2. Connect a ceramic capacitor between BST1 and LX1. See the

High-Side Gate-Driver Supply (BST_) section.

27

28

BST1

DH1

High-Side Gate-Driver Output for Buck 2. DH1 output voltage swings from V

to V

.

LX1

BST1

Exposed Pad. Connect EP to ground. Connecting the exposed pad to ground does not remove the

requirement for proper ground connections to PGND1, PGND2, and AGND. The exposed pad is

attached with epoxy to the substrate of the die, making it an excellent path to remove heat from the

IC.

—

EP

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Undervoltage Lockout (UVLO)

Detailed Description

The BIAS input undervoltage-lockout (UVLO) circuitry

inhibits switching if the 5V bias supply (BIAS) is below its

2.6V (typ) UVLO falling threshold. Once BIAS rises above

its UVLO rising threshold and EN1 and EN2 enable the

buck controllers, the controllers start switching and the

output voltages begin to ramp up using soft-start.

The MAX20034 is an automotive-rated dual-out-

put switching power-supply IC. The IC integrates two

synchronous step-down controllers and can provide two

independent-controlled power rails as follows:

● Buck controller with a fixed 5V output voltage, or an

adjustable 1V to 10V output voltage.

Buck Controllers

● Buck controller with a fixed 3.3V output voltage, or an

adjustable 1V to 10V output voltage.

The IC provides two buck controllers with synchronous

rectification. The step-down controllers use a pulse-width

modulation (PWM) current-mode control scheme. External

MOSFETs allow for optimized load-current design. Fixed-

frequency operation with optimal interleaving minimizes

input ripple current from the minimum to the maximum

input voltages. Output-current sensing provides an accu-

rate current limit with a sense resistor, or power dissipation

can be reduced using lossless current sensing across the

inductor.

EN1 and EN2 enable the respective buck controllers.

Connect EN1 and EN2 directly to V

supply sequencing logic.

, or to power-

BAT

In skip mode, the total supply current is reduced to 17μA

(typ) with Buck 1 disabled and Buck 2 enabled. When

both controllers are disabled, the total current drawn is

further reduced to 6.5µA (typ).

Fixed 5V Linear Regulator (BIAS)

The internal circuitry of the IC requires a 5V bias supply.

An internal 5V linear regulator (BIAS) generates this bias

supply. Bypass BIAS with a ≥ 6.8µF ceramic capacitor to

guarantee stability under the full-load condition.

Soft-Start

Once a buck converter is enabled by driving the corre-

sponding EN_ high, the soft-start circuitry gradually ramps

up the reference voltage during soft-start time (t

SSTART

The internal linear regulator can source up to 100mA

(150mA under EXTVCC switchover; see the EXTVCC

Switchover section). Use the following equation to

estimate the internal current requirements for the IC:

= 5ms (typ)) to reduce the input surge currents during

startup. Before the IC can begin the soft-start, the follow-

ing conditions must be met:

1) V

exceeds the 3.25V (max) undervoltage-lockout

BIAS

I

= I

+ f

(Q

+ Q

+

threshold.

BIAS

CC

SW

G_DH1

G_DL1

Q

+ Q

) = 10mA to 50mA (typ)

G_DH2

G_DL2

2) V

is logic-high.

EN_

where I

is the internal 5mA (typ) supply current, f

SW

CC

Switching Frequency/External Synchronization

is the switching frequency, and Q

is the MOSFET’s

G_

The IC provides an internal oscillator, adjustable from

220kHz to 2.2MHz. High-frequency operation optimizes

the application for the smallest component size, trading off

efficiency to higher switching losses. Low-frequency opera-

tion offers the best overall efficiency at the expense of

component size and board space. To set the switching fre-

total gate charge (specification limits at V

minimize the internal power dissipation, bypass BIAS to

an external 5V rail.

= 5V). To

GS

EXTVCC Switchover

The internal linear regulator can be bypassed by connect-

ing an external supply (3.25V to 5.2V) or one of the buck

converter outputs to EXTVCC. BIAS internally switches to

EXTVCC and the internal linear regulator turns off. This

configuration has several advantages:

quency, connect a resistor (R ) from FOSC to AGND:

FOSC





R_FOSC

6





25.5 +

















f_SW =

● It reduces the internal power dissipation of the device.

R_FOSC

● The low-load efficiency improves as the internal supply

current is scaled down proportionally to the duty cycle.

See the Typical Operating Characteristics to determine

the relationship between switching frequency and R

.

If V

drops below 3.25V, the internal regulator is

FOSC

EXTVCC

enabled and BIAS switches back to the internal regulator.

The IC can be synchronized to an external clock by con-

necting the external clock signal to FSYNC. A rising edge

on FSYNC resets the internal clock. The FSYNC clock

should have a minimum 150ns high pulse width.

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Light-Load Eﬃciency Skip Mode (V

= 0V)

MOSFET Gate Drivers (DH_ and DL_)

FSYNC

Drive FSYNC low to enable skip mode. In skip mode, the

IC stops switching until the FB_ voltage drops below the

reference voltage. Once the FB_ voltage has dropped

below the reference voltage, the IC begins switching until

the inductor current reaches 30% (skip threshold) of the

maximum current defined by the inductor DCR or output

shunt resistor.

The DH_ high-side n-channel MOSFET drivers are

powered from capacitors at BST_, while the low-side

drivers (DL_) are powered by the 5V linear regulator

(BIAS). On each channel, a shoot-through protection

circuit monitors the gate-to-source voltage of the external

MOSFETs to prevent a MOSFET from turning on until the

complementary switch is fully off. There must be a low-

resistance, low-inductance path from the DL_ and DH_

drivers to the MOSFET gates for the protection circuits to

work properly. Follow the instructions listed to provide the

necessary low-resistance and low-inductance path:

Forced-PWM Mode (V

= High)

FSYNC

Driving FSYNC high prevents the IC from entering skip

mode by disabling the zero-crossing detection of the induc-

tor current. This forces the low-side gate-driver waveform

to constantly be the complement of the high-side gate-

driver waveform, so the inductor current reverses at light

loads and discharges the output capacitor. The benefit

of forced-PWM (FPWM) mode is to keep the switching

frequency constant under all load conditions; however,

forced-frequency operation diverts a considerable amount

of the output current to PGND, reducing the efficiency

under light-load conditions.

● Use very short, wide traces (50 mils to 100 mils wide

if the MOSFET is 1in from the driver).

● It may be necessary to decrease the slew rate for the

gate drivers to reduce switching noise or to compen-

sate for low-gate-charge capacitors. For the low-side

drivers, use 1nF to 5nF gate capacitors from DL_ to

PGND, and for the high-side drivers, connect a small

5Ω to 10Ω resistor between BST_ and the bootstrap

capacitor.

FPWM mode is useful for improving load-transient

response and eliminating unknown frequency harmonics

that can interfere with AM radio bands.

Note: Gate drivers must be protected during shutdown,

at the absence of the supply voltage (V

= 0V) when

BIAS

the gate is pulled high either capacitively or by the leak-

age path on the PCB; therefore, external-gate pulldown

resistors are needed to prevent making a direct path from

Maximum Duty-Cycle Operation

The IC has a maximum duty cycle of 97% (min). The inter-

nal logic of the IC looks for approximately 10 consecutive

high-side FET-on pulses and decides to turn on the low-

side FET for 150ns (typ) every 12μs. The input voltage at

which the IC enters dropout changes depending on the

input voltage, output voltage, switching frequency, load

current, and the efficiency of the design. The input voltage

at which the IC enters dropout can be approximated as:

V

BAT

to PGND.

High-Side Gate-Driver Supply (BST_)

The high-side MOSFET is turned on by closing an inter-

nal switch between BST_ and DH_ and transferring the

bootstrap capacitor’s (at BST_) charge to the gate of the

high-side MOSFET. This charge refreshes when the high-

side MOSFET turns off and the LX_ voltage drops down to

ground potential, taking the negative terminal of the capaci-

tor to the same potential. At this time, the bootstrap diode

recharges the positive terminal of the bootstrap capacitor.

V

OUT_

= [V

+ (I

x R

)]/0.97

OUT_

ON_H

Note: The above equation does not take into account

the efficiency and switching frequency, but is a good first-

order approximation. Use the R

the data sheet of the high-side MOSFET used.

(max) number from

ON_H

The selected n-channel high-side MOSFET determines the

appropriate boost capacitance values (C

in the Typical

BST_

Operating Circuit) according to the following equation:

Spread Spectrum

Q

G

The IC features enhanced EMI performance, which per-

forms ±6% dithering of the switching frequency to reduce

peak emission noise at the clock frequency and its har-

monics, making it easier to meet stringent emission limits.

C

=

BST_

∆V

BST_

where Q is the total gate charge of the high-side

G

MOSFET and ΔV

on the high-side MOSFET driver after turn-on. Choose

is the voltage variation allowed

BST_

When using an external clock source (i.e., driving the

FSYNC input with an external clock), spread spectrum is

disabled.

ΔV such that the available gate-drive voltage is not

BST_

significantly degraded (e.g., ΔV

= 100mV to 300mV)

BST_

when determining C

.

BST_

The boost capacitor should be a low-ESR ceramic

capacitor. A minimum value of 100nF works in most cases.

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Current Limiting and

Overcurrent Protection

Current-Sense Inputs (OUT_ and CS_)

If the inductor current in the IC exceeds the maximum

current limit programmed at CS_ and OUT_, the respec-

tive driver turns off. In an overcurrent mode, this results in

shorter and shorter high-side pulses.

The current-limit circuit uses differential current-sense

inputs (OUT_ and CS_) to limit the peak inductor current.

If the magnitude of the current-sense signal exceeds the

current-limit threshold (V

= 80mV (typ)), the PWM

LIMIT1,2

A hard short results in a minimum on-time pulse every

clock cycle. Choose the components so they can with-

stand the short-circuit current if required.

controller turns off the high-side MOSFET. The actual

maximum load current is less than the peak current-

limit threshold by an amount equal to half the inductor

ripple current; therefore, the maximum load capability is

a function of the current-sense resistance, inductor value,

Overvoltage Protection

The IC limits the output voltage of the buck converters

by turning off the high-side gate driver at approximately

109% of the regulated output voltage. The output voltage

needs to come back in regulation before the high-side

gate driver starts switching again.

switching frequency, and duty cycle (V

/V ).

OUT_ IN

For the most accurate current sensing, use a current-

sense shunt resistor (R ) between the inductor and the

SH

output capacitor. Connect CS_ to the inductor side of R

SH

and OUT_ to the capacitor side. Dimension R such that

SH

LOAD,MAX

Design Procedure

the maximum inductor current (I

= I

+ 1/2

across R

L,MAX

I ) induces a voltage of V

RIPPLE,PP

,

SH

LIMIT1,2

Buck Converter Design Procedure

Eﬀective Input Voltage Range in Buck Converters

Although the MAX20034 can operate from input supplies

up to 36V (42V transients) and regulate down to 1V, the

including all tolerances.

For higher efficiency, the current can also be measured

directly across the inductor. This method could cause

up to 30% error over the entire temperature range and

requires a filter network in the current-sense circuit. See

the Current-Sense Measurement section.

minimum voltage conversion ratio (V

/V ) might be

OUT IN

limited by the minimum controllable on-time. For proper

fixed-frequency PWM operation and optimal efficiency,

Buck 1 and Buck 2 should operate in continuous conduc-

tion during normal operating conditions. For continuous

conduction, set the voltage-conversion ratio as follows:

Voltage Monitoring (PGOOD_)

The IC includes several power-monitoring signals to facili-

tate power-supply sequencing and supervision. PGOOD_

can be used to enable circuits that are supplied by the cor-

responding voltage rail, or to turn on subsequent supplies.

V

OUT

> t

× f

ON(MIN)

SW

V

IN

Each PGOOD_ goes high (high impedance) when the cor-

responding regulator output voltage is in regulation. Each

PGOOD_ goes low when the corresponding regulator

output voltage drops below 92% (typ) or rises above 95%

(typ) of its nominal regulated voltage. Connect a 10kΩ (typ)

pullup resistor from PGOOD_ to the relevant logic rail to

level-shift the signal.

where t

is 50ns (typ) and f

is the switching

ON(MIN)

SW

frequency in Hz. If the desired voltage conversion does not

meet the above condition, pulse skipping occurs to

decrease the effective duty cycle. Decrease the switching

frequency if constant switching frequency is required. The

same is true for the maximum voltage conversion ratio.

PGOOD_ asserts low during soft-start, soft-discharge, and

when either buck converter is disabled (either EN1 or EN2

is low).

The maximum voltage conversion ratio is limited by the

maximum duty cycle (97%).

V

OUT

< 0.97

Thermal-Overload, Overcurrent, and

Overvoltage/Undervoltage Behavior

V

− V

DROP

IN

where V

= I

(R

+ R

) is the sum of

DCR

Thermal-Overload Protection

DROP

OUT

ON,HS

the parasitic voltage drops in the high-side path, and f

is the programmed switching frequency. During low-drop

operation, the device reduces f to 80kHz. In practice,

the above condition should be met with adequate margin

SW

Thermal-overload protection limits total power dissipation in

the IC. When the junction temperature exceeds +170°C, an

internal thermal sensor shuts down the device, allowing it to

cool. The thermal sensor turns the device on again after the

junction temperature cools by 20°C.

SW

for good load-transient response.

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

The ratio of the inductor peak-to-peak AC current to DC

average current (LIR) must be selected first. A good initial

value is a 30% peak-to-peak ripple current to average-

current ratio (LIR = 0.3). The switching frequency, input

voltage, output voltage, and selected LIR then determine

the inductor value as follows:

Setting the Output Voltage in Buck Converters

Connect FB1 and FB2 to BIAS to enable the fixed buck

controller output voltages (5V and 3.3V) set by a preset

internal resistive voltage-divider connected between

the output (OUT_) and AGND. To externally adjust the

output voltage between 1V and 10V, connect a resistive

divider from the output (OUT_) to FB_ to AGND (see

(V − V

) x D

OUT

IN

the Typical Operating Circuit). Calculate R

and

L

=

MIN1

FB_1

f

x I

x LIR

R

with the following equation:

SW OUT

, and I are typical values (so that

OUT

FB_2

where V , V

IN

OUT









V

OUT_

efficiency is optimum for typical conditions).





FB_2



 −1

R

= R

FB_1





V





FB_





The next equation ensures that the inductor current

downslope is less than the internal slope compensation:





where V

table).

= 1V (typ) (see the Electrical Characteristics

FB_

m2

m ≥

DC output accuracy specifications in the Electrical

Characteristics table refer to the error comparator’s thresh-

2

V

OUT

L

m2 =

× A

× R

VCS CS

old, V

= 1V (typ). When the inductor conducts continu-

FB_

ously, the device regulates the peak of the output ripple,

so the actual DC output voltage is lower than the slope-

compensated trip level by 50% of the output ripple voltage.

where m is the internal slope compensation, m2 is the

inductor current downslope, A is current-sense gain

VCS

In discontinuous-conduction mode (skip or STDBY active

(11V/V), and R

is current-sense resistor.

CS

and I

< I ), the device regulates the valley of

LOAD(SKIP)

OUT

Solving for L and adding 1.5 multiplier to account for

tolerances in the system:

the output ripple, so the output voltage has a DC regulation

level higher than the error-comparator threshold.

Inductor Selection in Buck Converters

Three key inductor parameters must be specified

for operation with the MAX20034: inductance value (L),

( V

× A

× R

×1.5)

CS

OUT

VCS

L

=

MIN2

2m

inductor saturation current (I

), and DC resistance

SAT

Select the larger of L

and L

as L .

MIN

MIN1

MIN2

(R

). To determine the optimum inductance, knowing

DCR

the typical duty cycle (D) is important.

The maximum inductor value is recommended to:

= 1.6 x L

L

MAX

MIN

V

OUT

D =

or D =

For example, in a typical-use case, 5V output voltage and

5A output current, R is 15mΩ, m is 0.4V/µs for 2.2MHz

V

−I

(R

+ R )

DCR

IN

IN OUT DS(ON)

CS

if the R

of the inductor and R

of the MOSFET

and 0.08V/µs for 400kHz. For 2.2MHz:

DCR

DS(ON)

are available with V = (V

- V ). All values should

DIODE

IN

BAT

be typical to optimize the design for normal operation.

(5×11× 0.015×1.5)

L

=

= 1.5µH

MIN

2× 0.4

Inductance

(

)

The exact inductor value is not critical and can be adjust-

ed to make trade-offs among size, cost, efficiency, and

transient-response requirements.

L

= 1.6×1.5 = 2.4µH

MAX

Therefore, a 2.2µH inductor is chosen for 2.2MHz.

For 400kHz, L is calculated as 7.7µH and L

● Lower inductor values increase LIR, which minimizes

size and cost, and improves transient response at the

cost of reduced efficiency due to higher peak currents.

is

MIN

MAX

12.3µH; therefore, a 10µH inductor is chosen.

● Higher inductance values decrease LIR, which

increases efficiency by reducing the RMS current at

the cost of requiring larger output capacitors to meet

load-transient specifications.

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Current-Sense Measurement

Peak Inductor Current

For the best current-sense accuracy and overcurrent

protection, use a ±1% tolerance current-sense resistor

between the inductor and output, as shown in Figure

1A. This configuration constantly monitors the inductor

current, allowing accurate current-limit protection. Use low-

inductance current-sense resistors for accurate measure-

ment.

Inductors are rated for maximum saturation current. The

maximum inductor current equals the maximum load

current, in addition to half the peak-to-peak ripple

current:

∆I

INDUCTOR

I

= I

+

LOAD(MAX)

PEAK

2

Alternatively, high-power applications that do not require

highly accurate current-limit protection can reduce the

overall power dissipation by connecting a series RC circuit

across the inductor (see Figure 1B) with an equivalent time

constant:

For the selected inductance value, the actual peak-to-peak

inductor ripple current (ΔI ) is calculated as:

INDUCTOR

V

(V − V

)

OUT IN

OUT

x L

SW

∆I

=

INDUCTOR

V

x f

IN

R2





where ΔI

is in mA, L is in µH, and f

is in kHz.

R

=

R

DCR

INDUCTOR

SW

CSHL









R1+ R2

Once the peak current and inductance are known, the

inductor can be selected. The saturation current should

and:

be larger than I

, or at least in a range where the

PEAK

L

1





inductance does not degrade significantly. The MOSFETs

are required to handle the same range of current without

dissipating too much power.

R

=

+

DCR









C

R1 R2

EQ

where R

is the required current-sense resistor and

CSHL

MOSFET Selection in Buck Converters

R

is the inductor’s series DC resistor. Use the

DCR

Each step-down controller drives two external logic-level

n-channel MOSFETs as the circuit switch elements. The

key selection parameters to choose these MOSFETs

include the items in the following sections.

inductance and R

values provided by the inductor

DCR

manufacturer. If DCR sense is the preferred current-

sense method, the recommended resistor value for R1

(Figure 1B) should be less than 1kΩ.

Threshold Voltage

All four n-channel MOSFETs must be a logic-level type,

with guaranteed on-resistance specifications at V

4.5V. If the internal regulator is bypassed (for example:

Carefully observe the PCB layout guidelines to ensure

the noise and DC errors do no corrupt the differential

current-sense signals seen by CS_ and OUT_. Place the

sense resistor close to the device with short, direct traces,

making a Kelvin-sense connection to the current-sense

resistor.

=

GS

V

= 3.3V), the n-channel MOSFETs should be

EXTVCC

chosen to have guaranteed on-resistance at that gate-to-

source voltage.

Input Capacitor in Buck Converters

Maximum Drain-to-Source Voltage (V

DS(MAX)

All MOSFETs must be chosen with an appropriate V

rating to handle all V voltage conditions.

)

The discontinuous input current of the buck converters

cause large input ripple currents and therefore the input

capacitor must be carefully chosen to withstand the input

ripple current and keep the input-voltage ripple within

design requirements. The 180° ripple phase operation

increases the frequency of the input-capacitor ripple

current to twice the individual converter switching

frequency. When using ripple phasing, the worst-case

input-capacitor ripple current is when the converter with

the highest output current is on.

DS

IN

Current Capability

The n-channel MOSFETs must deliver the average

current to the load and the peak current during switching.

Choose MOSFETs with the appropriate average current

at V

= 4.5V, or V

= V

when the internal

GS

EXTVCC

linear regulator is bypassed. For load currents below

approximately 3A, dual MOSFETs in a single package

can be an economical solution. To reduce switching noise

for smaller MOSFETs, use a series resistor in the BST_

path and additional gate capacitance. Contact the factory

for guidance using gate resistors.

The input-voltage ripple is composed of ΔV (caused by

Q

the capacitor discharge) and ΔV

(caused by the ESR

ESR

of the input capacitor). The total voltage ripple is the sum

of ΔV and ΔV that peaks at the end of an on-cycle.

Q

ESR

Maxim Integrated

│ 16

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

INPUT (V )

IN

C

IN

MAX20034

NH

NL

DH_

LX_

DL_

R

L

SENSE

C

OUT

GND

CS_

OUT_

A) OUTPUT SERIES-RESISTOR SENSING

INPUT (V )

IN

C

IN

MAX20034

NH

INDUCTOR

DH_

L

DCR

LX_

C

OUT

R1

R2

NL

DL_

R2

C

EQ

R

=

R

DCR

CSHL

GND

(_{R1 + R2})

L

1

R2

+

CS_

R

=

DCR

[_R1]

C

EQ

OUT_

B) LOSSLESS INDUCTOR SENSING

Figure 1. Current-Sense Configurations

Maxim Integrated

│ 17

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Calculate the input capacitance and ESR required for a

specific ripple using the following equation:

The amount of overshoot (V

no-load transient due to stored inductor energy can be

calculated as:

) during a full-load to

SOAR

∆V

ESR

2

ESR[Ω] =

(∆I

) L

LOAD(MAX)

∆I





P− P

2

V

≈

SOAR

I

+





LOAD(MAX)

2C

V

OUT OUT













V

ESR Considerations

OUT

V

I

x



LOAD(MAX)

The output-filter capacitor must have low enough equiva-

lent series resistance (ESR) to meet output ripple and

load-transient requirements, yet have high enough ESR

to satisfy stability requirements. When using high-capac-

itance, low-ESR capacitors, the filter capacitor’s ESR

dominates the output-voltage ripple, so the output capaci-

tor’s size depends on the maximum ESR required to meet

IN 

C

[µF] =

IN

∆V x f

(

)

Q

SW

where:

V

− V

x V

OUT

(

=

)

IN

OUT

x f

∆I

P−P

V

x L

IN SW

the output-voltage ripple (V

) specifications:

x LIR

LOAD(MAX)

RIPPLE(P-P)

I

is the maximum output current in A, ΔI

is

P-P

LOAD(MAX)

V

= ESR x I

the peak-to-peak inductor current in A, f

is the switch-

RIPPLE(P−P)

SW

ing frequency in MHz, and L is the inductor value in µH.

In standby mode, the inductor current becomes

discontinuous, with peak currents set by the idle-mode

The internal 5V linear regulator (BIAS) includes an output

UVLO with hysteresis to avoid unintentional chattering

during turn-on. Use additional bulk capacitance if the

input source impedance is high. At lower input voltage,

additional input capacitance helps avoid possible under-

shoot below the undervoltage-lockout threshold during

transient loading.

current-sense threshold (V

= 26mV (typ)).

CS,SKIP

Transient Considerations

The output capacitor must be large enough to absorb

the inductor energy while transitioning from no-load to

full-load condition without tripping the overvoltage fault

protection. The total output voltage sag is the sum of the

voltage sag while the inductor is ramping up, and the volt-

age sag before the next pulse can occur. Therefore:

Output Capacitor in Buck Converters

The actual capacitance value required relates to the

physical size needed to achieve low ESR, as well as to

the chemistry of the capacitor technology. The capacitor

is usually selected by ESR and the voltage rating rather

than by capacitance value.

2

L ∆I

(

)

LOAD(MAX)

C

=

OUT

2V

(V x D

− V

)

OUT

SAG IN

MAX

When using low-capacity filter capacitors, such as

ceramic capacitors, size is usually determined by the

∆I

t − ∆t

(

)

LOAD(MAX)

+

V

SAG

capacity needed to prevent V

and V

from

SAG

SOAR

causing problems during load transients. Generally, once

enough capacitance is added to meet the overshoot

requirement, undershoot at the rising load edge is no

longer a problem (see the Transient Considerations

section). However, low-capacity filter capacitors typically

have high-ESR zeros that can affect the overall stability.

where D

is the maximum duty factor (approximately

MAX

95%), L is the inductor value in µH, C

capacitor value in µF, t is the switching period (1/f ) in

µs, and Δt equals (V

is the output

OUT

SW

/V ) x t.

OUT IN

The IC uses a peak current-mode-control scheme that

regulates the output voltage by forcing the required

current through the external inductor, so the controller

uses the voltage drop across the DC resistance of the

inductor or the alternate series current-sense resistor

to measure the inductor current. Current-mode control

eliminates the double pole in the feedback loop caused

by the inductor and output capacitor, resulting in a smaller

phase shift and requiring less elaborate error-amplifier

The total voltage sag (V

) can be calculated as follows:

SAG

L(∆I

2

)

LOAD(MAX)

((V ×D

V

=

SAG

2C

) − V

)

OUT

IN

MAX

∆I

(t − ∆t)

LOAD(MAX)

+

C

OUT

Maxim Integrated

│ 18

www.maximintegrated.com

MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

compensation than voltage-mode control. A single series

resistor (R ) and capacitor (C ) is all that is required

(typ), and R

is the output resistance of the error

OUT,EA

amplifier, which is 2.2MΩ (typ) (see the Electrical

Characteristics table.)

C

to have a stable, high-bandwidth loop in applications

where ceramic capacitors are used for output filter-

ing (see Figure 2). For other types of capacitors, due

to the higher capacitance and ESR, the frequency of

the zero created by the capacitance and ESR is lower

than the desired closed-loop crossover frequency. To

stabilize a nonceramic output capacitor loop, add another

A dominant pole (f

capacitor (C ) and the amplifier output resistance

) is set by the compensation

dpEA

C

(R

). A zero (f

OUT,EA

) is set by the compensation

ZEA

resistor (R ) and the compensation capacitor (C ). There

C

is an optional pole (f

output capacitor ESR zero if it occurs near the crossover

) set by C and R to cancel the

PEA

F C

compensation capacitor (C ) from COMP to AGND to

F

frequency (f , where the loop gain equals 1 (0dB)). Thus:

C

cancel this ESR zero.

1

The basic regulator loop is modeled as a power

modulator, output feedback-divider, and an error amplifier,

as shown in Figure 2. The power modulator has a DC

f

=

dpEA

2π × C × (R

+ R )

C

OUT,EA

1

gain set by g

x R , with a pole and zero pair set

LOAD

f

=

mc

zEA

pEA

2π × C × R

C

by R , the output capacitor (C ), and its ESR. The

LOAD OUT

loop response is set by the following equations:

1

GAIN = g ×R

f

=

MOD(dc)

mc

LOAD

2π × C × R

F

where R

= V

/I

in Ω and g =1/(A

LOAD

OUT LOUT(MAX) mc V_CS

The loop-gain crossover frequency (f ) should be set

C

below 1/15th the switching frequency and much higher

x R ) in S. A

is the voltage gain of the current-sense

DC

V_CS

amplifier and is typically 11V/V. R

the inductor or the current-sense resistor in Ω.

is the DC resistance of

DC

than the power-modulator pole (f ). Select a value

pMOD

for f in the range:

C

In a current-mode step-down converter, the output

capacitor and the load resistance introduce a pole at the

following frequency:

f

SW

15

f

<< f

≤

C

pMOD

1

At the crossover frequency, the total loop gain must be

equal to 1. Therefore:

f

=

pMOD

2π × C

× R

LOAD

OUT

The unity-gain frequency of the power stage is set by

and g

V

FB

C

:

OUT

mc

GAIN

×

× GAIN

= 1

)

C

MOD(f

)

EA(f

C

V

OUT

g_mc

2π × C

f

=

UGAINpMOD

OUT

GAIN

= g

×R

EA(f

)

m,EA

C

The output capacitor and its ESR also introduce a zero at:

1

f

pMOD

GAIN

= GAIN

×

f

=

MOD(f

)

MOD(dc)

zMOD

C

f

2π × ESR× C

C

OUT

Therefore:

GAIN

Solving for R :

When C

is composed of “n” identical capacitors in

OUT

V

FB

×

parallel, the resulting C

= n x C

, and ESR

OUT

OUT(EACH)

× GAIN

= 1

)

EA(f

C

V

= ESR

/n. Note that the capacitor zero for a parallel

(EACH)

OUT

combination of alike capacitors is the same as for an

individual capacitor.

C

V

The feedback voltage-divider has a gain of GAIN

=

FB

OUT

R

=

C

V

/V

, where V is 1V (typ).

g

× V ×GAIN

FB MOD(f

FB OUT FB

m,EA

)

C

The transconductance error amplifier has a DC gain

of GAIN = g x R , where g is

EA(DC)

m,EA

OUT,EA

m,EA

the error-amplifier transconductance, which is 470µS

Maxim Integrated

│ 19

www.maximintegrated.com

MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Set the error-amplifier compensation zero formed by R

C

and C at the f

. Calculate the value of C as follows:

C

pMOD

C

g

= 1/(A

x R

)

DC

VCS

mc

1

CS_

C

=

C

CURRENT-MODE

POWER

2π × f

× R

C

pMOD

OUT_

MODULATION

If f

is less than 5 x f , add a second capacitor (C )

C F

zMOD

from COMP to AGND. The value of C is:

F

R1

R2

g

= 470µS

mea

R

ESR

1

FB_

C

=

F

2π × f

× R

C

COMP_

ERROR

AMP

zMOD

C

OUT

As the load current decreases, the modulator pole also

decreases; however, the modulator gain increases accord-

ingly and the crossover frequency remains the same.

R

V

REF

C

2.2MΩ

C

F

C

The following is a numerical example to calculate the

compensation network component values of Figure 2:

● A

= 11V/V

V_CS

DCR

Figure 2. Compensation Network

● R

= 15mΩ

● g = 1/(A

x R ) = 1/(11 x 0.015) = 6.06

DC

mc

V_CS

Applications Information

● V

= 5V

OUT

Layout Recommendations

● I

= 5.33A

OUT(MAX)

Careful PCB layout is critical to achieve low switching

losses and clean, stable operation. The switching power

stage requires particular attention (see Figure 3). If pos-

sible, mount all the power components on the top side of

the board, with their ground terminals flush against one

another. Follow these guidelines for good PCB layout:

● R

● C

= V

/I

= 5V/5.33A = 0.9375Ω

LOAD

OUT OUT(MAX)

= 2x47µF = 94µF

OUT

● ESR = 9mΩ/2 = 4.5mΩ

● f = 0.403MHz

SW

● Keep the high-current paths short, especially at the

ground terminals. This practice is essential for stable,

jitter-free operation.

GAIN

= 6.06×0.9375 = 5.68

MOD(dc)

1

f

=

≈ 1.8kHz

pMOD

● Keep the power traces and load connections short.

This practice is essential for high efficiency. Using

thick copper PCBs (2oz vs. 1oz) can enhance full load

efficiency by 1% or more.

2π × 94µF× 0.9375

f

SW

15

f

<< f

≤

C

pMOD

● Minimize current-sensing errors by connecting CS_ and

1.8kHz << f ≤ 80.6kHz

C

OUT_. Use kelvin sensing directly across the current-

sense resistor (R

). A high-frequency filter is

SENSE_

select f = 25kHz:

C

required if operating above 1.8MHz. The recommended

RC filter values are 20Ω/100pF. Refer to the MAX20034

EV kit data sheet schematic for details.

1

f

=

≈ 376kHz

zMOD

2π × 4.5mΩ × 94µF

● Route high-speed switching nodes (BST_, LX_, DH_,

and DL_) away from sensitive analog areas (FB_, CS_,

and OUT_).

since f

> f :

C

zMOD

● R ≈ 25kΩ

C

● C ≈ 3.3nF

C

● C ≈ 18pF

F

Maxim Integrated

│ 20

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

from DH_ to the gate of the external HS switch and

from the LX_ pin to the inductor. Up to 100mA of

current flow from the BIAS capacitor through the

bootstrap diode to the bootstrap capacitor. Dimension

those traces accordingly.

Layout Procedure

1) Place the power components first, with ground

terminals adjacent (low-side FET, C , C , and

IN

OUT_

Schottky). If possible, make all these connections on

the top layer with wide, copper-filled areas.

4) Make the DC-DC controller ground connections as

shown in Figure 3. This diagram can be viewed as

having two separate ground planes: power ground,

where all the high-power components go; and an

analog ground plane for sensitive analog components.

The analog ground plane and power ground plane

must meet only at a single point directly under the IC.

2) Mount the controller IC adjacent to the low-side

MOSFET, preferably on the back side opposite DL_

and DH_ to keep LX_, PGND, DH_, and the DL_ gate

drive lines short and wide. The DL_ and DH_ gate

traces must be short and wide (50 mils to 100 mils

wide if the MOSFET is 1in from the controller IC) to

keep the driver impedance low and for proper adaptive

dead-time sensing.

5) Connect the output-power planes directly to the output-

filter capacitor positive and negative terminals with

multiple vias. Place the entire DC-DC converter circuit

as close as possible to the load.

3) Group the gate-drive components (BST_ diode and

capacitor and LDO bypass capacitor BIAS) together

near the controller IC. Be aware that gate currents of

up to 1A flow from the bootstrap capacitor to BST_,

KELVIN-SENSE VIAS

UNDER THE SENSE RESISTOR

(REFER TO THE MAX20034 EVALUATION KIT)

INDUCTOR

LOW-SIDE

n-CHANNEL

MOSFET (NH)

C

OUT

OUTPUT

GROUND

HIGH-SIDE

n-CHANNEL

MOSFET (NL)

INPUT

Figure 3. Layout Example

Maxim Integrated

│ 21

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MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Typical Operating Circuit

R

PGOOD1

FB1

PGOOD1

DL1

CS1

COUT1

OUT1

CS1

OUT1

CS1

R

CS1

*

OUT1

LX1

L1

BST1

EN1

MAX20034

DH1

PGND2

FSYNC

FOSC

COMP1

AGND

COMP2

EN2

BIAS

PGND1

EXTVCC

OUT1

V

BAT

IN

C

IN

DH2

BST2

LX2

R

*

CS2

L2

OUT2

CS2

OUT2

C

OUT2

CS2

DL2

PGOOD2

FB2

PGND2

*DCR SENSE IS ALSO AN OPTION.

Maxim Integrated

│ 22

www.maximintegrated.com

MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Ordering Information

V

ADJUSTABLE

1V to 10V

OUT

SPREAD

SPECTRUM

PART

TEMP RANGE

PIN-PACKAGE

FIXED

5V/3.3V

MAX20034ATIR/VY+

MAX20034ATIS/VY+

-40°C to +125°C

28 TQFN-EP**

Oﬀ

1V to 10V

On

/V denotes an automotive qualified part.

+Denotes a lead(Pb)-free/RoHS-compliant package.

**EP = Exposed pad.

Chip Information

PROCESS: BiCMOS

Maxim Integrated

│ 23

www.maximintegrated.com

MAX20034

Automotive High-Eﬃciency 2.2MHz, 36V, Dual Buck

Controller with 17μA Quiescent Current

Revision History

REVISION REVISION

PAGES

DESCRIPTION

CHANGED

NUMBER

DATE

0

1

9/17

Initial release

Removed future product status from MAX20034ATIR/VY+ in Ordering Information

—

2/18

23

Updated the Simpliﬁed Block Diagram, TOC01–TOC02, TOC08, TOC11–TOC12,

TOC23, Pin Description table, and the Inductance and Transient Considerations section.

2, 6–7, 9, 11,

15, 19–20

2

3

4/18

8/19

Updated title to indicate automotive part; updated Beneﬁts and Features, TOC5 inTypi-

cal Operating Characteristics, and Pin Description

1, 6, 10

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.

Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses

are implied. Maxim Integrated reserves the right to change the circuitry and speciﬁcations without notice at any time. The parametric values (min and max limits)

shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.

©

Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.

2018 Maxim Integrated Products, Inc.

│ 24

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