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™

QPROX QT310

LQ

PROGRAMMABLE

CAPACITANCE

SENSOR IC

ꢀ Single channel digital advanced capacitance sensor IC

ꢀ Spread spectrum burst modulation for high EMI rejection

ꢀ Full autocal capability

ꢀ User programmable via cloning process

ꢀ Internal eeprom storage of user setups, cal data

ꢀ Variable drift compensation & recalibration times

ꢀ BG and OBJ cal modes for learn-by-example

ꢀ Sync pins for daisy-chaining or noise suppression

ꢀ Variable gain via Cs capacitor change

ꢀ Selectable output polarity, high or low

ꢀ Toggle mode (optional via setups)

ꢀ Push-pull output

ꢀ Completely programmable output behavior

via cloning process from a PC

ꢀ HeartBeat™ health indicator (can be disabled)

APPLICATIONS

ꢀ Fluid level sensors

ꢀ Industrial panels

ꢀ Appliance controls

ꢀ Security systems

ꢀ Access controls ꢀ Material detection

ꢀ Micro-switch replacement ꢀ Toys & games

This device requires only a few external passive parts to operate. It uses spread-spectrum burst modulation to dramatically

reduce interference problems.

The QT310 charge-transfer (“QT’”) touch sensor IC is a self-contained digital IC capable of detecting proximity, touch, or fluid

level when connected to a corresponding type of electrode. It projects sense fields through almost any dielectric, like glass,

plastic, stone, ceramic, and wood. It can also turn metal-bearing objects into intrinsic sensors, making them respond to

proximity or touch. This capability coupled with its ability to self calibrate continuously or to have fixed calibration by example

can lead to entirely new product concepts.

It is designed specifically for advanced human interfaces like control panels and appliances or anywhere a mechanical switch

or button may be found; it can also be used for material sensing and control applications, and for point-level fluid sensing.

The ability to daisy-chain permits electrodes from two or more QT310’s to be adjacent to each other without interference. The

burst rate can be programmed to a wide variety of settings, allowing the designer to trade off power consumption for response

time.

The IC’s RISC core employs signal processing techniques pioneered by Quantum; these are specifically designed to make

the device survive real-world challenges, such as ‘stuck sensor’ conditions and signal drift. All operating parameters can be

user-altered via Quantum’s cloning process to alter sensitivity, drift compensation rate, max on-duration, output polarity,

calibration mode, Heartbeat™ feature, and toggle mode. The settings are permanently stored in onboard eeprom.

The Quantum-pioneered HeartBeat™ signal is also included, allowing a host controller to monitor the health of the QT310

continuously if desired.

By using Quantum’s advanced, patented charge transfer principle, the QT310 delivers a level of performance clearly superior

to older technologies yet is highly cost-effective.

AVAILABLE OPTIONS

T_A

0⁰C to +70⁰C

-40⁰C to +85⁰C

SOIC

-

QT310-IS

8-PIN DIP

QT310-D

-

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QT310/R1.03 21.09.03

lowers susceptibility to EMI, and yet permits excellent

response time. Internally the signals are digitally processed to

reject impulse noise, using a 'consensus' filter which requires

several consecutive confirmations of a detection before the

output is activated.

Table 1-1 Pin Descriptions

1

Name

/CAL_CLR

/SYNC_O

SNS1

Function

Ext Cal, latch clear input

Sync Output

2

3

Sense 1 line

A unique cloning process allows the internal eeprom of the

device to be programmed to permit unique combinations of

sensing and processing functions.

4

VSS

Negative supply (ground)

Sense 2 line

Sync Input

5

SNS2

/SYNC_I

OUT

6

7

Detection output

Positive supply

+2 to 5 Vdc

100nF

8

VDD

Alternate Pin Functions for Cloning

10K 10K

3

6

7

SCK

SDO

SDI

Serial clone data clock

Serial clone data out

Serial clone data in

Calibration

1

6

7

2

3

5

/CAL

SYNC_O

SNS1

SYNC_I

OUT

ELECTRODE

C_s

C_x

SNS2

4.7nF

1 - OVERVIEW

The QT310 is a digital burst mode charge-transfer (QT)

sensor designed for touch controls, level sensing and

proximity sensing; it includes all hardware and signal

processing functions necessary to provide stable sensing

under a wide variety of changing conditions. Only one low

cost sampling capacitor is required for operation.

VSS

4

A unique aspect of the QT310 is the ability of the designer to

‘clone’ a wide range of user-defined setups into the part’s

eeprom during development and in production. Cloned setups

can dramatically alter the behavior of the part. For production,

the parts can be cloned in-circuit or can be procured from

Quantum pre-cloned.

Figure 1-1 Basic QT310 circuit

1.2 ELECTRODE DRIVE

1.2.1 SWITCHING

O

PERATION

Figure 1-1 shows the basic QT310 circuit using the device,

with a conventional output drive and power supply

connections.

The IC implements direct-to-digital capacitance acquisition

using the charge-transfer method, in a process that is better

understood as a capacitance-to-digital converter (CDC). The

QT switches and charge measurement functions are all

internal to the IC (Figure 1-2).

1.1 BASIC OPERATION

The QT310 employs bursts of charge-transfer cycles to

acquire its signal. Burst mode permits power consumption in

the microamp range, dramatically reduces RF emissions,

The CDC treats sampling capacitor Cs as a floating store of

accumulated charge which is switched between the sense

pins; as a result, the sense electrode can be connected to

either pin with no performance difference. In both cases the

rule Cs >> Cx must be observed for proper operation. The

polarity of the charge build-up across Cs during a burst is the

same in either case. Typical values of Cs range from 10nF to

200nF.

Result

SNS1

Larger values of Cx cause charge to be transferred into Cs

more rapidly, reducing available resolution and resulting in

lower gain. Conversely, larger values of Cs reduce the rise of

differential voltage across it, increasing available resolution

and raising gain. The value of Cs can thus be increased to

allow larger values of Cx to be tolerated (Figures 5-1 to 5-2).

Cs

Cx

Start

Done

As Cx increases, the length of the burst decreases resulting in

lower signal numbers.

SNS2

The electrode should always be connected to SNS1;

connections to SNS2 are also possible but this can cause the

signal to be susceptible to noise.

Charge

Amp

It is important to limit the amount of stray Cx capacitance on

both SNS terminals, especially if the Cx load is already large.

Figure 1-2 Internal Switching

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This can be accomplished by minimising trace lengths and

widths.

1.3.2 KIRCHOFF

’

S

C

URRENT

L

AW

Like all capacitance sensors, the QT310 relies on Kirchoff’s

Current Law (Figure 1-4) to detect the change in capacitance

of the electrode. This law as applied to capacitive sensing

requires that the sensor’s field current must complete a loop,

returning back to its source in order for capacitance to be

sensed. Although most designers relate to Kirchoff’s law with

regard to hardwired circuits, it applies equally to capacitive

field flows. By implication it requires that the signal ground

and the target object must both be coupled together in some

manner for a capacitive sensor to operate properly. Note that

there is no need to provide actual hardwired ground

1.2.2 CONNECTION TO

E

LECTRODE

The PCB traces, wiring, and any components associated with

or in contact with SNS1 and SNS2 will become touch

sensitive and should be treated with caution to limit the touch

area to the desired location.

Multiple electrodes can be connected, for example to create a

control button on both sides of an object, however it is

impossible for the sensor to distinguish between the two

electrodes.

connections; capacitive coupling to ground (Cx1) is always

sufficient, even if the coupling might seem very tenuous. For

example, powering the sensor via an isolated transformer will

provide ample ground coupling, since there is capacitance

between the windings and/or the transformer core, and from

the power wiring itself directly to 'local earth'. Even when

battery powered, just the physical size of the PCB and the

object into which the electronics is embedded will generally

be enough to couple a few picofarads back to local earth.

The implications of Kirchoff’s law can be most visibly

demonstrated by observing the E3B eval board’s sensitivity

change between laying the board on a table versus holding

the board in your hand by it’s batteries. The effect can also be

observed by holding the board by the electrode ‘Sensor1’,

letting it recalibrate, then touching the battery end; the board

will work quite well in this mode.

Figure 1-3 Mesh Electrode Geometry

1.3.3 VIRTUAL

C

APACITIVE

G

ROUNDS

When detecting human contact (e.g. a fingertip), grounding of

the person is never required, nor is it necessary to touch an

exposed metal electrode. The human body naturally has

several hundred picofarads of ‘free space’ capacitance to the

local environment (Cx3 in Figure 1-4), which is more than two

orders of magnitude greater than that required to create a

return path to the QT310 via earth. The QT310's PCB

however can be physically quite small, so there may be little

‘free space’ coupling (Cx1 in Figure 1-4) between it and the

environment to complete the return path. If the QT310 circuit

ground cannot be grounded via the supply connections, then

a ‘virtual capacitive ground’ may be required to increase

return coupling.

1.2.3 BURST

M

ODE

O

PERATION

The acquisition process occurs in bursts (Figure 1-7) of

variable length, in accordance with the single-slope CDC

method. The burst length depends on the values of Cs and

Cx. Longer burst lengths result in higher gains and more

sensitivity for a given threshold setting, but consume more

average power and are slower.

Burst mode operation acts to lower average power while

providing a great deal of signal averaging inherent in the CDC

process, making the signal acquisition process more robust.

The QT method is a very low impedance method of sensing

as it loads Cx directly into a very large capacitor (Cs). This

results in very low levels of RF susceptibility.

1.3 ELECTRODE DESIGN

1.3.1 ELECTRODE

G

EOMETRY AND

S

IZE

There is no restriction on the shape of the electrode; in most

cases common sense and a little experimentation can result

in a good electrode design. The QT310 will operate equally

well with a long, thin electrode as with a round or square one;

even random shapes are acceptable. The electrode can also

be a 3-dimensional surface or object. Sensitivity is related to

electrode surface area, orientation with respect to the object

being sensed, object composition, and the ground coupling

quality of both the sensor circuit and the sensed object.

Smaller electrodes have less sensitivity than large ones.

If a relatively large electrode surfaces are desired, and if tests

show that an electrode has a high Cx capacitance that

reduces the sensitivity or prevents proper operation, the

electrode can be made into a mesh (Figure 1-3) which will

have a lower Cx than a solid electrode area.

Figure 1-4 Kirchoff’s Current Law

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of the QT310, sensitivity can be high enough (depending on

Cx and Cs) that 'walk-by' signals are a concern; if this is a

problem, then some form of rear shielding may be required.

1.4 SENSITIVITY ADJUSTMENTS

There are three variables which influence sensitivity:

1. Cs (sampling capacitor)

2. Cx (unknown capacitance)

3. Signal threshold value

There is also a sensitivity dependence of the whole device on

Vdd. Cs and Cx effects are covered in Section 1.2.1.

The threshold setting can be adjusted independently from 1 to

255 counts of signal swing (Section 2.3).

Note that sensitivity is also a function of other things like

electrode size, shape, and orientation, the composition and

aspect of the object to be sensed, the thickness and

composition of any overlaying panel material, and the degree

of mutual coupling of the sensor circuit and the object (usually

via the local environment, or an actual galvanic connection).

Threshold levels of less than 5 counts in BG mode are not

advised; if this is the case, raise Cs so that the threshold can

also be increased.

Figure 1-5 Shielding Against Fringe Fields

1.4.1 INCREASING

S

ENSITIVITY

In some cases it may be desirable to greatly increase

sensitivity, for example when using the sensor with very thick

panels having a low dielectric constant, or when sensing low

capacitance objects.

A ‘virtual capacitive ground’ can be created by connecting the

QT310’s own circuit ground to:

(1) A nearby piece of metal or metallized housing;

(2) A floating conductive ground plane;

(3) A fastener to a supporting structure;

(4) A larger electronic device (to which its output might be

connected anyway).

Sensitivity can be increased by using a bigger electrode,

reducing panel thickness, or altering panel composition.

Increasing electrode size can have diminishing returns, as

high values of Cx load will also reduce sensor gain (Figures

5-1 and 5-2). The value of Cs also has a dramatic effect on

sensitivity, and this can be increased in value up to a limit.

Because the QT310 operates at a relatively low frequency,

about 500kHz, even long inductive wiring back to ground will

usually work fine.

Increasing electrode surface area will not substantially

increase sensitivity if its area is already larger than the object

to be detected. The panel or other intervening material can be

made thinner, but again there are diminishing rewards for

Free-floating ground planes such as metal foils should

maximise exposed surface area in a flat plane if possible. A

square of metal foil will have little effect if it is rolled up or

crumpled into a ball. Virtual ground planes are more effective

and can be made smaller if they are physically bonded to

other surfaces, for example a wall or floor.

1.3.4 FIELD

S

HAPING

The electrode can be prevented from sensing in undesired

directions with the assistance of metal shielding connected to

circuit ground (Figure 1-5). For example, on flat surfaces, the

field can spread laterally and create a larger touch area than

desired. To stop field spreading, it is only necessary to

surround the touch electrode on all sides with a ring of metal

connected to circuit ground; the ring can be on the same or

opposite side from the electrode. The ring will kill field

spreading from that point outwards.

If one side of the panel to which the electrode is fixed has

moving traffic near it, these objects can cause inadvertent

detections. This is called ‘walk-by’ and is caused by the fact

that the fields radiate from either surface of the electrode

equally well. Again, shielding in the form of a metal sheet or

foil connected to circuit ground will prevent walk-by; putting a

small air gap between the grounded shield and the electrode

will keep the value of Cx lower and is encouraged. In the case

Figure 1-6 Burst Detail

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QT310/R1.03 21.09.03

Figure 1-8 Burst when SC is set to 0 (no sleep cycles)

(Observed using a 750K resistor in series with probe)

Figure 1-7 Burst when SC is set to 1

(Observed using a 750K resistor in series with probe)

doing so. Panel material can also be changed to one having a The number of pulses in a burst and hence its duration

higher dielectric constant, which will help propagate the field.

Locally adding some conductive material to the panel

(conductive materials essentially have an infinite dielectric

constant) will also help; for example, adding carbon or metal

fibers to a plastic panel will greatly increase frontal field

strength, even if the fiber density is too low to make the

plastic electrically conductive.

increases with Cs and decreases with Cx.

1.5.2 BURST

S

PACING: TBS, TSC

Between acquisition bursts, the device can go into a low

power sleep mode. The duration of this is a multiple of Tsc,

the basic sleep cycle time. Tsc depends heavily on Vdd as

shown in Figure 5-4, page 16. The parameter SC calls out

how many of these cycles are used. More SC means lower

power but also slower response time.

1.4.2 DECREASING

S

ENSITIVITY

In some cases the circuit may be too sensitive, even with high

signal threshold values. In this case gain can be lowered by

making the electrode smaller, using sparse mesh with a high

space-to-conductor ratio (Figure 1-3), and most importantly by

decreasing Cs. Adding Cx capacitance will also decrease

sensitivity.

Tbs is the spacing from the start of one burst to the start of

the next. This timing depends on the burst length Tbd and the

dead time between bursts, i.e. Tsc.

The resulting timing of Tbs is:

Tbs = Tbd + (SC x Tsc)

-or-

Tbs = Tbd + 2.25ms

where SC > 0

where SC = 0

It is also possible to reduce sensitivity by making a capacitive

divider with Cx by adding a low-value capacitor in series with

the electrode wire.

If SC = 0, the device never sleeps between bursts (example:

Figure 1-8). In this case the value of Tsc is fixed at about

2.25ms, but this time is not spent in Sleep mode and maximal

power is consumed.

1.5 TIMING

Figure 1-7 and 1-8 shows the basic timing parameters of the

QT310. The basic QT310 timing parameters are:

if SC >> 0 (example: SC=15), the device will spend most of its

time in sleep mode and will consume very little power, but it

will be much slower to respond.

Tbd

Tbs

Tsc

Tmod

Tdet

Burst duration

Burst spacing

Sleep Cycle duration

Max On-Duration

Detection response time

(1.5.1)

(1.5.2)

(1.5.3)

(1.5.4)

By selecting a supply voltage and a value for SC, it is possible

to fine-tune the circuit for the desired speed / power trade-off.

1.5.3 MAX

O -DURATION, TMOD

N

1.5.1 BURST

F

REQUENCY AND

D

URATION

The Max On-Duration is the amount of time required for

sensor to recalibrate itself when continuously detecting. This

parameter is user-settable by changing MOD and SC (see

Section 2.6).

The burst duration depends on the values of Cs and Cx, and

to a lesser extend, Vdd. The burst is normally composed of

hundreds of charge-transfer cycles (Figure 1-6) operating at

about 240kHz. This frequency varies by about 7ꢀ during the

burst in a spread-spectrum modulation pattern. See Section

3.5.2 page 13 for more information on spread-spectrum.

Tmod restarts if the sensor becomes inactive before the end

of the Max On Duration period.

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QT310/R1.03 21.09.03

are crowded together with a rep rate that depends entirely on

the burst lengths (Section 1.5.1).

1.5.4 RESPONSE

Response time Tdet from the onset of detection to the OUT

pin becoming active depends on:

TIME, TDET

Response time, drift compensation rate, max on-duration, and

power consumption are all affected by this parameter. A high

value of SC will allow the device to consume very low power

but it will also be very slow.

Tbs Burst spacing (Section 1.5.2)

DIT Detection Integrator Target (user setting)

DIS Detect Integration Speed

Tbd Burst duration

(user setting)

(if DIS is set to ‘fast’)

2.2 DRIFT COMPENSATION (PDC, NDC)

Signal drift can occur because of changes in Cx, Cs, Vdd,

electrode contamination and ageing effects. It is important to

compensate for drift, otherwise false detections and sensitivity

shifts can occur.

If the control bit DIS is normal (0), then Tdet depends on the

rate at which the bursts are acquiring, and the value of DIT. A

DIT number of bursts must confirm the detection before the

OUT line becomes active:

Tdet = Tbs x DIT (normal DIS)

Drift compensation is performed by making the signal’s

reference level slowly track the raw signal while no detection

is in effect. The rate of adjustment must be performed slowly,

otherwise legitimate detections could be affected. The device

compensates using a slew-rate limited change to the signal

reference level; the threshold and hysteresis points are slaved

to this reference.

If DIS is set to ‘fast’, then Tdet is computed as:

Tdet = (SC x Tsc) + (DIT x (Tbd + 2.25ms)) (fast DIS)

Quantum’s QT3View software calculates an estimate of

response time based on this formula.

1.6 EXTERNAL RECALIBRATION

Once an object is detected, drift compensation stops since a

legitimate signal should not cause the reference to change.

The /CAL_CLR pin can be used to recalibrate the sensor on

demand. A low pulse of at least Tbs (burst spacing) duration

is require to initiate a recalibration. The calibration occurs just

after /CAL_CLR returns high.

Positive and negative drift compensation rates (PDC, NDC)

can be set to different values (Figure 2-1). This is invaluable

for permitting a more rapid reference recovery after the device

has recalibrated while an object was present and then

removed.

In BG1 mode (Section 2.8.4), the calibration data is not stored

in EEPROM, and the part will recalibrate after each power up.

In BG1 mode, if the device has been set for Toggle Latch

output mode, the /CAL_CLR pin becomes an output reset

control and the part cannot be recalibrated via /CAL_CLR.

However the part can be recalibrated by powering it down and

back up again (Section 2.7.3).

Positive drift occurs when the Cx slowly increases. Negative

drift occurs when Cx slowly decreases (see Section 2.8.1).

PDC+1 sets the number of burst spacings, Tbs, that

determines the interval of drift compensation, where:

Tbs = Tbd + (SC x Tsc)

where SC > 0 (Section 1.5.2)

-or-

In BG2 mode, the calibration data is stored in EEPROM, and

the part will not recalibrate after power up, using instead the

stored calibration data. The internal eeprom has a life

expectancy of 100,000 erase/write cycles.

Tbs = Tbd + 2.25ms

where SC = 0 (Section 1.5.2)

In OBJ mode, the part stores the calibration data into

EEPROM and the part will not recalibrate after power up,

using instead the stored calibration data.

Example: PDC = 9,

Tbs = 100ms

then

(user setting)

In both BG2 and OBJ mode, the device must be calibrated

using the /CAL_CLR input, or the calibration data can be set

via cloning process, otherwise the calibration data will be

invalid.

Tpdc = (9+1) x 100ms = 1 sec

NDC operates in exactly the same way as PDC.

2 - Control & Processing

All acquisition functions are digitally controlled and

can be altered via the cloning process.

Signals are processed using 16 bit integers, using

Quantum-pioneered algorithms specifically

designed to provide for high survivability.

2.1 SLEEP CYCLES (SC)

Range: 0..255; Default: 1

Affects speed & power of entire device.

Refer to Section 1.5.2 for more information on the

effect of Sleep Cycles.

SC changes the number of intervals Tsc

separating two consecutive burst (Figure 1-7 and

1-8). SC = 0 disables sleep intervals and bursts

Figure 2-1 Drift Compensation

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Hysteresis should be set to between 10ꢀ and 40ꢀ of the

threshold value for best results.

2.2.1 NEGATIVE

Range: 0..255; Default: 2; 255 disables

Compensation for drift with decreasing Cx

D

RIFT

COMPENSATION (NDC)

If HYS is set to 0, there will be no hysteresis (0ꢀ).

NDC corrects the reference when the internal signal is drifting

up, i.e. Cx is decreasing (see Section 2.8.1). Every interval of

time the device checks for the need to move its reference

level in the positive internal direction (negative Cx direction) in

accordance with signal drift. The resulting timing interval for

this adjustment is Tndc.

If THR = 10 and HYS = 2, the hysteresis zone will represent

20ꢀ of the threshold level. In this example the ‘hysteresis

zone’ is the region from 8 to 10 counts of signal level. Only

when the signal falls back to 7 will the OUT pin become

inactive.

This should normally be faster than positive drift

compensation in order to compensate quickly for the removal

of a touch or obstruction from the electrode after a MOD

recalibration (Section 1.5.3).

2.5 DETECT INTEGRATORS (DIA, DIB, DIS)

DIAT

DIBT

DIS

Range: 1..256 Default: 10

Range: 0, 1

Default: 1

Affects response time Tdet.

Use NDC+1 to compute actual drift timings.

See Figure 2-2 for operation.

2.2.2 POSITIVE

Range: 0...255 Default: 100; 255 disables

Compensation for drift with increasing Cx

D

RIFT

COMPENSATION (PDC)

It is usually desirable to suppress detections generated by

sporadic electrical noise or from quick contact with an object.

To accomplish this, the QT310 incorporates a pair of

detection integrator (‘DI’) counters that serve to filter out

This corrects the reference when the signal drifting down, i.e.

Cx is increasing (see Section 2.8.1). Every interval of time the sporadic noise. These counters can also have the effect of

device checks for the need to move its reference level in the

negative internal direction (positive Cx direction) in

accordance with signal drift. The resulting timing interval for

this adjustment is Tpdc.

slowing down response time if desired.

DI counters act as a powerful noise filter.

These DI counters work with spread-spectrum modulation to

drastically suppress the effects of external RFI. See page 13

for details.

This value should not be set too fast, since an approaching

finger could be compensated for partially or entirely before

even touching the sense electrode.

DIA / DIAT: The first counter, DIA, increments after each

burst if the signal threshold has been exceeded, until DIA

reaches its terminal count DIAT, after which the OUT pin is

activated. If the signal falls below the threshold level prior to

reaching DIAT, DIA is immediately reset to zero.

Use PDC+1 to compute actual drift timings.

2.3 THRESHOLD (THR)

Range: 1..255; Default: 6

Affects sensitivity; not used in OBJ mode.

DIA can also be viewed as a 'consensus' filter that requires

signal threshold crossings over ‘T’ successive bursts to create

an output, where ‘T’ is the terminal count (DIAT).

The detection threshold is measured in terms of counts of

signal deviation with respect to the reference level. Higher

threshold counts equate to less sensitivity since the signal

must travel further in order to cross the detection point.

DIB / DIBT: If OUT has been active and the signal falls below

the hysteresis level, a second detection integrator, DIB,

counts up.

If the signal equals or exceeds the threshold value, a

detection can occur. The detection will end only when the

signal become less than the hysteresis

level.

When DIBT is reached, OUT is deactivated.

THR is not used in OBJ mode (Section

2.8.5). In OBJ mode the threshold is set by

example during calibration.

2.4 HYSTERESIS (HYS)

Range: 0...255; Default: 2; 0 disables

Affects detection stability.

Hysteresis is measured in terms of counts

of signal deviation relative to the threshold

level. Higher values equate to more

hysteresis. The device will become inactive

after a detection when the Cx level moves

below THR-HYS in normal mode or above

THR+HYS in absence mode (Section2.8.2)

Hysteresis helps prevents chattering of the

OUT pin.

If HYS is set to a value equal or greater than

THR, the device may malfunction.

Figure 2-2 Detect Integrators Operation (Section 2.5)

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QT310/R1.03 21.09.03

DISA / DISB: Because the DI counters count at the burst rate,

slow burst spacings can result in very long detection delays

with terminal counts above 1. To cure this problem, the burst

rate can be made faster while DIA or DIB are counting. This

creates the effect of a gear-shifted detection process: normal

speed when there are no threshold crossings, and fast mode

when a detection is pending.

2.7 OUTPUT FEATURES

Available output processing options accommodate most

requirements; these can be set via the clone process.

If TOG and TOGL modes are disabled, OUT responds to

detections with a steady-state active logic level which lasts for

the duration of a detection, until a MOD timeout occurs

(Section 2.6).

DISA and DISB respectively gearshift the effect of DIA and

DIB. The gear-shifting ceases and normal speed resumes

once the detection is confirmed (DIA = DIAT) and once the

detection ceases (DIB = DIBT).

The OUT pin is push-pull CMOS.

2.7.1 POLARITY (OUTP)

Options: active-low or -high; Default: active-low

When SC=0 the device operates without any sleep cycles,

and so the timebase for the DI counters is very fast.

The polarity of OUT can be set via option OUTP using the

cloning process. Either active-low or active-high can be

selected. This not the same as ‘direction of signal detection’

(Section 2.8.1).

2.6 MAX ON-DURATION (MOD)

Range: 0..255; Default: 14; 255 disables

Affects parameter Tmod, the calibration delay time

In ‘active high’ mode the normal, inactive polarity of OUT is

low; in ‘active low’ mode the normal, inactive polarity of OUT

is high.

If a stray object remains on or near the sense electrode, the

signal may rise enough to activate the OUT pin thus

preventing normal operation. To provide a way around this, a

Max On-Duration (‘MOD’) timer is provided to cause a

recalibration if the activation lasts longer than the designated

timeout, Tmod.

OUTP also selects the initial state of OUT when the sensor is

used in Toggle or Toggle Latch modes (Sections 2.7.2, 2.7.3);

for example, if OUTP is set active-low, the initial state of OUT

after power-up will be high.

The MOD function can also be disabled, in which case the

sensor will never recalibrate unless the part is powered down

and back up again. In infinite timeout the designer should take

care to ensure that drift in Cs, Cx, and Vdd do not cause the

device to ‘stick on’ inadvertently when the target object is

removed from the sense field.

2.7.2 TOGGLE

MODE (TOG)

Options: enabled or disabled; Default: disabled

Toggle mode gives the OUT pin a touch-on / touch-off flip-flop

action, so that its state changes with each new detection. It is

most useful for controlling power loads, for example kitchen

appliances, power tools, light switches, etc.

MOD is expressed in multiples of the burst space interval,

which can be either Tbs or Tbd depending on the Sleep

Cycles setting (SC).

MOD time-outs (Section 2.6) and the /CAL_CLR pin will

recalibrate the sensor but leave the OUT state unchanged.

If SC > 0, the delay is:

The OUTP option (Section 2.7.1) sets the initial state of the

sensor after power-up.

Tmod = (MOD + 1) x 16 x Tbs

Example:

2.7.3 TOGGLE

L

ATCH

MODE (TOGL)

Options: enabled or disabled; Default: disabled

Tbs = 100ms,

MOD = 9;

In this mode, OUT becomes active when a valid detection

occurs but will only go inactive again if an external clear signal

is applied to the part; further detections after the first one will

not change the state of OUT.

Tmod = (9 + 1) x 16 x 100ms = 160 secs.

If SC = 0, Tmod is a function of the total combined burst

durations, Tbd. If SC = 0, the delay is:

Tmod = (MOD + 1) x 256 x Tbd

The external clear signal is applied to the /CAL_CLR pin

which functions only as latch clear input if TOGL is enabled.

The only way to recalibrate the sensor externally in TOGL

mode is to cycle power off and back on.

Example:

Tbd = 18ms,

MOD = 9;

A logic low pulse on /CAL_CLR will clear the latch and make

OUT inactive. As the /CAL_CLR pin is sampled once per

burst, the clear pulse has to be at least as long as Tbs (the

burst duration) to ensure the latch clears.

Tmod = (9 + 1) x 256 x 18ms = 46 secs.

If MOD = 255, recalibration timeout = infinite (disabled)

regardless of SC.

An MOD induced recalibration will make the OUT pin inactive

except if the output is set to toggle mode (Section 2.7.2), in

which case the OUT state will be unaffected but the sensor

will have recalibrated.

If any underlying threshold detection remains active for longer

than the Max On-Duration (MOD) period the device will

recalibrate automatically, but the OUT pin will not change

state.

A clear pulse applied to /CAL_CLR will clear the latch even if

the part is in the process of recalibrating due to a MOD

timeout.

The clear state of OUT can be set via the OUTP option

(Section 2.7.1).

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QT310/R1.03 21.09.03

Toggle Latch Mode cannot be used with BG2 or OBJ modes,

as /CAL_CLR must be used as a calibrate input in these two

modes (Sections 1.6, 2.8.4, 2.8.5).

2.8.2 SENSE

DIRECTION (SD)

OPTIONS: POS OR

N

EG;

D

EFAULT

: POSITIVE

The programmable SD option controls whether the device

responds to increases in Cx (‘normal’ detection) or decreases

in Cx (‘absence’ detection). The default mode is positive.

2.7.4 HEART

B

EAT™ OUTPUT (HB)

Default: Enabled

Setup: Enable/Disable;

The OUT pin can have HeartBeat™ ‘health’ indicator pulses

superimposed on it. This operates by floating the 'OUT' pin for

approximately 15µs before each burst.

2.8.2.1 Positive Sense Direction (default)

This is the normal mode of operation for touch sensing.

Calibration is normally done when an object is not present;

OUT becomes active if an object approaches.

Heartbeat can be used to determine if the sensor is operating

properly. The frequency of the floats can be used to see if the

IC is operating within desired limits. The Heartbeat signal can

be tested by connecting a 10K resistor to OUT that is toggled

by a microcontroller depending on the logic level of OUT.

In this configuration, if Cx increases enough the internal

signal will pass the threshold level, and OUT will become

active. Cx must fall again so the internal signal traverses the

hysteresis level for OUT to become inactive.

Heartbeat pulses can be removed simply by placing a

capacitor on the OUT pin; if OUT is loaded into a high-

impedance CMOS input or MOSFET, this is usually enough.

The threshold and hysteresis levels are set relative to the

reference level determined during calibration.

2.8.2.2 Negative Sense Direction

It is possible to disable HeartBeat provided SC is set to zero,

by setting the HB control bit to '1'. Otherwise, the Heartbeat

signal is always enabled.

In this mode, if the part is made to calibrate when an object is

present, OUT will become active if the object departs (Cx

decreases).

In this configuration, if Cx decreases enough the internal

signal will pass the threshold level, and OUT will become

active. Cx must rise again so the internal signal traverses the

hysteresis level for OUT to become inactive.

2.7.5 OUTPUT

D

RIVE

C

The OUT pin is a push-pull CMOS type.

APABILITY

OUT can source or sink up to 2mA of non-inductive current. If

an inductive load is used, such as a small relay, the load

should be diode-clamped to prevent damage. The current

must be limited to 2mA max continuous to prevent detection

side effects from occurring, which happens when the load

current creates voltage drops on the die and bonding wires;

these small shifts can materially influence the signal level to

cause detection instability.

The threshold and hysteresis levels are set relative to the

reference level determined during calibration.

2.8.3 DETECT

M

ODE (DM) SELECTION

O

PTIONS: BG OR OBJ; EFAULT: BG

D

The IC can be set to calibrate and detect in one of two

different modes to suit the application. The selection is made

using the cloning process.

2.8 DETECTION MODES

SD - Sense Direction: Pos or Neg; Default: Positive

The device default is BG. There are two BG modes, BG1 and

BG2, which must be further selected as described below. The

BG mode default is BG1.

DM - Detect Mode: BG or OBJ;

BG - BG Mode: BG1 or BG2;

Default: BG

Default: BG1

It is possible to change the basic way the device detects and

operates via the cloning process as described below. In

particular, it is possible to determine whether the device

responds to increases in Cx (‘normal’ detection) or decreases

in Cx (‘absence’ detection). It is also possible to change how

the device calibrates itself, in one of three possible modes.

OBJ mode is described in Section 2.8.5.

2.8.4 BG (BACKGROUND) DETECTION

M

ODES

O

PTIONS: BG1 OR BG2; EFAULT: BG

D

1

The BG modes are useful when it is easier to calibrate on the

baseline signal level than the signal from the object to be

detected. The detection is always made relative to this

reference level, and the sensitivity is governed by the

adjustable threshold level (as well as capacitor Cs, and load

Cx). The BG modes are generally easier to use than OBJ.

2.8.1 SIGNAL

D

EFINITIONS

Increasing Cx load on the electrode will result in a shorter

burst length. Since internal computations are based on burst

length, a shorter burst length means a smaller internal signal

number; conversely, a longer burst length means less Cx but

higher internal signal numbers. In summary:

There are two BG modes, BG1 and BG2. In these modes,

threshold and hysteresis values are calculated relative to the

reference level, which in turn is determined during calibration.

The two modes differ in that BG1 mode the calibration is

volatile whereas in BG2 mode the calibration reference is

stored in eeprom and reused until the next calibration.

Cx rises ꢀ shorter Burst Length ꢀ less internal signal

Cx drops ꢀ longer Burst Length ꢀ more internal signal

These relationships, are important to understand to avoid

confusion. They mirror signal values shown in QT3View and

the burst length as viewed on an oscilloscope.

Hysteresis can be altered as per Section 2.4.

Sense direction (SD) behavior: In both BG modes OUT can

be made active on either positive or negative Cx changes

(Section 2.8.2). SD selection affects which side of the

reference the threshold and hysteresis points are placed.

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QT310/R1.03 21.09.03

In addition, the OUT pin can be made either active low or

active high (Section 2.7.1).

2.8.5 OBJ (OBJECT) DETECTION

M

ODE

This mode is useful to do a ‘learn by example’ calibration.

Typically, a test object is placed at the electrode in such a

way as to create a 50ꢀ signal level change relative to a

normal, full presentation of the object. The QT310 is then

calibrated in OBJ mode. Calibration in OBJ mode should

never be done with a full presentation of signal, as this will

create a marginal, unreliable detection.

2.8.4.1 BG1 Mode (volatile reference)

In BG1 mode, the reference is set via recalibration initiated

using the /CAL_CLR pin or on power-up. The resulting

reference level is not stored into EEPROM. Max On-Duration

and drift compensation are able to function normally.

BG1 mode is useful when the signal can change slightly over

time and temperature, and it is useful to track these changes

without a loss of sensitivity.

This mode is suited to material detection, fluid level sensing,

and similar applications.

In OBJ mode, on calibration the current signal value is

recorded as a fixed threshold point and stored to EEPROM.

2.8.4.2 BG2 Mode (stored reference)

In BG2 mode, the reference level is fixed and stored in

internal EEPROM. Drift compensation (Section 2.2) can be

used, but changes to the reference due to drift compensation

are not updated to EEPROM. Max On-Duration can also be

enabled (Section 2.6); if a MOD timeout occurs, the new

reference will be stored in EEPROM.

The hysteresis level is made relative to the fixed threshold,

and can be altered as with the BG modes. If hysteresis is too

large, the sensor can ‘stick’ on; hysteresis should normally be

set to a small value, just enough to prevent output chatter.

Hysteresis can also be made intentionally large, for example

for ‘bang-bang’ fluid level sensing, where an ‘upper’ level is

calibrated using OBJ, and a ‘lower’ cut-out level is defined by

the hysteresis value. The sensor must have SD = positive for

this mode (Section 2.8.2).

The reference is normally set during recalibration when the

/CAL_CLR pin pulses low (Section 1.6); the resulting

reference value is then stored in EEPROM. At power-up the

part automatically restores this reference level and runs

without another recalibration.

OBJ mode does not make use of a reference level and does

not allow drift compensation or Max On-Duration to operate.

The threshold point is fixed for all time until another

The reference value can also be entered numerically via the

cloning process (Table 4-1, page 14) to precisely replicate the /CAL_CLR signal is received.

calibration point across many devices.

The OBJ threshold value can also be entered numerically via

BG2 mode is useful when it is desired to lock in the reference the cloning process (Table 4-1, page 14) to precisely replicate

to prevent changes on startup, for example to replace

mechanical switches in process controls.

the threshold point across many devices.

Positive, negative detection mode behavior: In OBJ mode

OUT can be made active on either positive or negative signal

changes (Section 2.8.2). The signal direction selection affects

which side of the threshold the hysteresis level is placed after

calibration.

The OUT pin can be made either active low or active high

(Section 2.7.1).

U1

Vdd

1

6

2

7

3

5

OUT1

/CAL

OUT

Vdd

2.9 SYNCHRONISATION

Open Loop

The synchronization feature allows a QT310 to generate its

burst on demand from an external trigger rather than of its

own accord. This feature is made possible by the fact that the

QT310 operates in burst mode, rather than continuously.

Sync is a powerful feature that permits two important

operating modes: Daisy-chaining, and noise synchronization.

SENSOR 1

/SYNC_I SNS1

/SYNC_O SNS2

Closed Loop

CS1

U2

Vdd

1

6

2

7

3

5

OUT2

Daisy-chaining allows several QT310 or similar devices to

coexist in close proximity to each other without cross

interference. Noise synchronization allows a QT310 to lock

onto the fundamental frequency of an external interference

source, such as 50/60Hz, to correlate the noise with the

signal and thus eliminate alias frequencies from the acquired

signal. These are extremely powerful noise reduction

methods.

/CAL

OUT

SENSOR 2

/SYNC_I SNS1

/SYNC_O SNS2

CS2

Un

Vdd

1

6

2

7

3

5

OUT_N

/CAL

OUT

The SYNC_I pin is used to trigger the QT310 to generate a

burst. The sleep timer will always wake the part if a sync

pulse has not been received before the sleep time expires.

The sleep timer is always restarted when a sync pulse is

received.

SENSOR N

/SYNC_I SNS1

/SYNC_O SNS2

CS3

The pulse applied to SYNC_I must be normally high,

negative-going, and of >15µs pulse duration. SYNC_O emits

an 80µs pulse at the end of each burst.

Figure 2-3 Daisy chain wiring

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QT310/R1.03 21.09.03

During fast integration (Section 2.5), when bursts are

generated quickly a number of times in sequence without

regard to the sleep timer, a single SYNC_O pulse is

It is also possible to devise a tree structure of devices, where

some devices in the chain trigger two or more slaves. This

speeds up the acquisition process considerably, but some

generated only after the last burst in the series of fast spaced thought must be given to timing considerations so that

adjacent electrodes do not have bursts which overlap each

other in time.

bursts in order to prevent downstream slave parts from being

triggered too rapidly.

If SC=0 (no sleep cycles), no Sync_O pulses are generated.

After the burst has completed the QT310 checks the level on

SYNC_I. If SYNC_I is high, the part goes back to sleep; if

SYNC_I is still low the device waits until the SYNC_I is high

again before going back to sleep. If this is the case, power

drain will be higher so it is important to limit the pulse width to

an amount less than the burst length (but greater than

>15µs).

Disabling Sync: Connecting Sync_I to +Vdd will disable Sync

and the part will acquire bursts at the normal rate. If Sync is at

Vss, the device will wait for a Sync pulse, until the Tsc period

Vdd

2.9.2 NOISE

S

YNCHRONIZATION

U2:A

74HC14

C1

100pF

Using the sync feature, a QT310 can be synchronized to a

repetitive external source of interference such as the power

line frequency (Figure 2-4) in order to dramatically reduce

signal noise. If line frequency is present near the sensors, this

feature should be used.

R1

1M

R3

1M

R4

4.7k - 10K

R2

470K-1M

Line Input

2.2nF

C2

Vdd

U1

With this circuit the sensor can tolerate up to 100V/M of AC

electric field. It is particularly useful for line-powered touch

controls.

8

VDD

7

1

6

2

OUT1

OUT

/CAL

Noise sync and daisy-chaining can be combined by having

the first device in the chain sync to the external noise source.

3

5

SENSOR

SNS1

SNS2

/SYNC_I

CS

/SYNC_O

3 Circuit Guidelines

VSS

3.1 SAMPLE CAPACITORS

4

Cs capacitors can be virtually any plastic film or low to

medium-K ceramic capacitor. The normal usable Cs range is

from 10nF ~ 200nF depending on the sensitivity required;

larger values of Cs require higher stability to ensure reliable

sensing. Acceptable capacitor types include NP0 or C0G

ceramic, PPS film, Polypropylene film, and X7R ceramic in

that order.

Figure 2-4 Line sync circuit

expires; at that point the part will acquire regardless of the

absence of a Sync pulse.

2.9.1 DAISY-CHAINING QT310’

S

3.2 POWER SUPPLY

One use for synchronization is where two or more QT310’s in

close proximity to each other are synchronously daisy-

chained to avoid crosstalk (Figure 2-3).

3.2.1 STABILITY

The QT310 derives its internal references from the power

supply. Sensitivity shifts and timing changes will occur with

changes in Vdd, as often happens when additional power

supply loads are switched on or off via the Out pin.

One QT310 should be designated as the ‘Master’; this part

should have the shortest SC sleep time, while the

downstream parts which depend on the master and any

intermediary devices should have longer sleep time settings

than the master.

These supply shifts can induce detection ‘cycling’, whereby an

object is detected, the load is turned on, the supply sags, the

detection is no longer sensed, the load is turned off, the

supply rises and the object is reacquired, ad infinitum.

The parts can be chained in a loop (Fig 2-4 switch set to

‘closed loop’); in this configuration the master will generate a

new burst after the last slave has finished, making the scan

sequence of all devices the most time-efficient possible. If the

master doesn’t received a pulse before the sleep time has

elapsed it will generate a new burst. This mode is most useful

if there are a relatively small number of devices in the chain

and there is a need for fast response.

Detection ‘stiction’, the opposite effect, can occur if a load is

shed when the output is active and the signal swings are

small: the Out pin can remain stuck even if the detected

object is no longer near the electrode.

3.2.2 SUPPLY

R

EQUIREMENTS

In open-loop, the rep rate of acquisition is set purely by the

burst rate of the master. It is possible in this mode to have

very long chains of parts with relatively good response time.

The disadvantage of this mode is that it is possible for the

bursts of downstream slaves to overlap with upstream

devices, potentially causing interference if their electrodes are

in physical proximity to each other.

Vdd can range from 2.0 to 5.0 volts. If Setups programming is

required during operation, the minimum Vdd is 2.2V. Current

drain will vary depending on Vdd, the chosen sleep cycles,

and the burst lengths. Increasing Cx values will decrease

power drain since increasing Cx loads decrease burst length

(Figures 5-1 and 5-2).

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QT310/R1.03 21.09.03

If the power supply is shared with another electronic system,

care should be taken to assure that the supply is free of

spikes, sags, and surges. In BG1 mode the QT310 will track

slow changes in Vdd if drift compensation is enabled, but it

can be adversely affected by rapid voltage steps and spikes

at the millivolt level.

VDD

100nF

8

VDD

RE3

RE4

RE5

RE2

RE1

1

2

6

7

3

5

CAL

OUT

If desired, the supply can be regulated using a conventional

low current regulator, for example CMOS LDO regulators with

low quiescent currents, or standard 78Lxx-series 3-terminal

regulators.

SYNC_O SNS1

SYNC_I SNS2

SENSOR

CS

For proper operation a 100nF (0.1uF) ceramic bypass

capacitor must be used between Vdd and Vss; the bypass

cap should be placed very close to the Vdd and Vss pins.

VSS

4

Figure 3-1 ESD/EMC protection resistors

3.3 PCB LAYOUT

3.3.1 GROUND

P

LANES

dielectric properties, panel thickness, and rise time of the

ESD transients.

The use of ground planes around the device is encouraged

for noise reasons, but ground should not be coupled too close

to the sense pins in order to reduce Cx load. Likewise, the

traces leading from the sense pins to the electrode should not

be placed directly over a ground plane; rather, the ground

plane should be relieved by at least 3 times the width of the

sense traces directly under it, with periodic thin bridges over

the gap to provide ground continuity.

ESD protection can be enhanced with an added resistor RE1

(Figure 3-1). As the transfer time is ~833ns, the circuit can

tolerate values of RE1 which result in an RC timeconstant of

1/6th this amount or about 140ns. The ‘C’ of the RC is the Cx

load. Thus, for Cx= 20pF, the maximum of RE1 should be

6.8K ohms. Larger amounts of RE1 or Cx may result in

noticeably reduced gain.

3.3.2 CLONE

P

ORT

C

ONNECTOR

If a cloning connector is used, place this close to the QT310.

Placing the cloning connector far from the QT310 will increase

the load capacitance Cx of the sensor line SNS1 and

decrease sensitivity. Long distances on these lines can also

make the cloning process more susceptible to communication

errors from ringing and interference.

3.5 EMC ISSUES

Electromagnetic and electrostatic susceptibility are often a

problem with capacitive sensors. QT310 behavior under these

conditions can be improved by adding RE1 (Figure 3-1),

exactly as for ESD protection. The resistor should be placed

next to the chip.

If the SYNC_I input is used, a 10K ohm resistor should be

used to avoid conflicts with the cloning process (Figure 4-1).

This works because the inbound RC network formed by RE1

and Cs has a very low cut-off frequency which can be

computed by the formula:

Cloning can be designed for production by using pads (SMT

or through-hole) on the solder side which are connected to a

fixture via spring loaded ATE-style ‘pogo-pins’. This eliminates

the need for an actual connector to save cost.

1

Fc =

2✜ RCs

If R = 6.8K and Cs = 10nF, then Fc = 2,340 Hz.

Important Note: Since SCK is shared on the SNS1 pin, it is

possible that stray external fields can cause these devices to

enter into Clone mode accidentally. If long wiring or large

electrodes are used that could pick up interference, install a

470K resistor from SNS1 to ground to suppress pickup. If the

device enters clone mode accidentally, it may be necessary to

cycle power to recover the device.

This leads to very strong suppression of external field effects.

Nevertheless, it is always wise to reduce lead lengths by

placing the QT310 as close to the electrode as possible.

3.4 ESD ISSUES

VDD

In cases where the electrode is placed behind a dielectric

panel, the device will usually be well protected from static

discharge. However, even with a plastic or glass panel,

transients can still flow into the electrode via induction, or in

extreme cases, via dielectric breakdown. Porous materials

may allow a spark to tunnel right through the material; partially

conducting materials like 'pink poly' static dissipative plastics

will conduct the ESD right to the electrode. Panel seams can

permit discharges through edges or cracks.

100nF

8

VDD

1

2

6

7

3

5

/CAL

/SYNC_O

/SYNC_I

C/AL

SDI

SCK

OUT

SENSOR

S/YNC_O

SDO

CS

SNS2

Testing is required to reveal any problems. The QT310 has

internal diode protection which can absorb and protect the

device from most induced discharges, up to 20mA; the

usefulness of the internal clamping will depend on the

VSS

4

Figure 4-1 Clone interface wiring

LQ

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QT310/R1.03 21.09.03

Likewise, RF emissions are sharply curtailed by the use of

RE1, which bandwidth limits RF emissions based on the value

of RE1 and Cx, the electrode capacitance.

4 Parameter Cloning

The cloning process allows user-defined settings to be loaded

into internal eeprom, or read back out, for development and

production purposes.

3.5.1 LINE

CONDUCTED EMI

Line conducted EMI can be reduced by making sure the

power supply is properly bypassed to chassis ground. The

OUT line can also be paths for conducted EMI, and these can

be bypassed to circuit ground with an RC filter network. The

additional resistors RE2 through RE5 can also help with

conducted EMI.

The QTM300CA cloning board in conjunction with QT3View

software simplifies the cloning process greatly. The E3B eval

board has been designed with a connector to facilitate direct

connection with the QTM300CA. The QTM300CA in turn

connects to any PC with a serial port which can run QT3View

software (included with the QTM300CA and available free on

Quantum’s web site).

3.5.2 SPREAD-SPECTRUM

M

ODULATION

The connections required for cloning are shown in Figure 4-1.

Further information on the cloning process can be found in

the QTM300CA instruction guide. Section 3.3.2 above

discusses wiring issues associated with cloning.

The QT310 uses spread-spectrum burst modulation to

dramatically reduce susceptibility to external noise sources.

Spread-spectrum is implemented using frequency hopping

between four ‘channels’ centered around 240kHz. The

frequency of operation is altered with each successive burst;

the total frequency spread is approximately 7ꢀ.

The parameters which can be altered are shown in Table 4-1,

page 14.

It is possible for a host controller to read and change the

internal settings via the interface connections shown, but

doing so will disturb the sensing process even when data

transfers are not occurring. The additional capacitive loading

of the interface pins will contribute to Cx; also, noise on the

interface lines can cause erratic operation.

Spread-spectrum operates full-time and cannot be disabled.

If the DIAT (Detect Integrator terminal count) is set to DIAT=2,

then two different frequencies will be used to determine a

detection result. There is no way to control which two

frequencies are used, but they are guaranteed to be different.

If the DIAT (Detect Integrator) is set to 4 or higher, the

detection process will take advantage of all four possible

frequencies before confirming a result. All it takes is one

‘clear’ frequency for a false detection to be suppressed, since

a non-detection on one sample is enough to clear the DI

counter and abort a pending detection.

The internal eeprom has a life expectancy of 100,000

erase/write cycles.

A serial interface specification for the device can be obtained

by contacting Quantum.

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QT310/R1.03 21.09.03

5 Electrical specifications

5.1 ABSOLUTE MAXIMUM SPECIFICATIONS

Operating temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . as designated by suffix

Storage temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -65^OC to +150^OC

VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.5 to +6V

Max continuous pin current, any control or drive pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±±0mꢀ

Short circuit duration to ground, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite

Short circuit duration to V^DD, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite

Eeprom Setups max write cycles. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100,000

Voltage forced onto any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -1V to (Vdd + 0.5) Volts

5.2 RECOMMENDED OPERATING CONDITIONS

V

DD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +2.0 to 5V

DD min required for eeprom programming of Setups. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +2.2V

V

Short-term supply ripple+noise. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±5mV

Long-term supply stability. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±100mV

Cs value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10nF to 200nF

Cx value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to 100pF

5.3 AC SPECIFICATIONS

Vdd = 3.0, Ta = recommended operating range, Cs=100nF unless noted

Parameter

Description

Recalibration time

Min

Typ

7

Max

Units

ms

ns

Notes

Cs, Cx dependent

TRC

T

PC

Charge/transfer duration

Burst center frequency

Burst frequency modulation

Burst length

833

240

7

Fc

kHz

%

FD

TBL

0.5

15

25

ms

µs

Cs = 4.7nF to 200nF; Cx = 0

T

HB

SIP

SOP

Heartbeat pulse width

Input sync pulse

15

T

µs

T

Output sync pulse

80

µs

5.4 SIGNAL PROCESSING

Description

Min

Typ

Max

Units

counts

samples

ms/level

secs

Notes

Threshold differential

1

0

1

255

254

256

Hysteresis

Consensus filter length

Positive drift compensation rate

Negative drift compensation rate

Post-detection recalibration timer duration

-

<1

infinite

5.5 DC specifications

Vdd = 3.0V, Cs = 10nF, Cx = 5pF, Ta = recommended range, unless otherwise noted

Parameter

Description

Supply current

Min

2

Typ

Max

Units

µA

V/s

V

Notes

IDD

600

1,500

V

DDS

Supply turn-on slope

Input low voltage

100

Required for proper start-up

Vdd = 2.5 to 5.0V

V

IL

0.3 Vdd

0.4

V

IH

Input high voltage

Low output voltage

High output voltage

Load capacitance range

Acquisition resolution

Sensitivity range

0.6 Vdd

V

Vdd = 2.5 to 5.0V

V

OL

V

OUT, 2mA sink

VOH

Vdd-0.6

0

V

OUT, 1.5mA source

CX

100

16

7

pF

bits

fF

A

R

S

1,000

Ref Figs. 5-1, 5-2

LQ

15

QT310/R1.03 21.09.03

10.00

1.00

0.10

0.01

10.00

1.00

0.10

0.01

4.7nF

9nF

4.7nF

9nF

19nF

43nF

74nF

124nF

200nF

19nF

43nF

74nF

124nF

200nF

0

10

20

30

40

50

0

10

20

30

40

50

Cx Load

Figure 5-2 Typical sensitivity vs Cx;

Threshold = 6, Vdd = 3.0 Volts

Figure 5-1 Typical sensitivity vs Cx;

Threshold = 16, Vdd = 3.0 Volts

180

160

140

120

100

80

25.000

20.000

15.000

10.000

60

5.000

0.000

40

Cx = 0pF

20

Cx = 21pF

52

0

118

228

507

Cx = 48pF

Load (pf)

884

1.5

2

2.5

3

3.5

4

4.5

5

5.5

1450

2357

Sampling Capacitor (nF)

Power Supply (Volts)

Figure 5-3 Typical Burst length vs Cx, Cs;

Vdd = 3.0 Volts

Figure 5-4 Tsc vs Vdd; SC = 1

LQ

16

QT310/R1.03 21.09.03

16

14

12

10

8

6

4

2

0

1.5

2

2.5

3

3.5

4

4.5

5

5.5

-2

Vdd (Volts)

Figure 5-5 Typical internal signal count change vs Vdd

5.00%

4.00%

3.00%

2.00%

1.00%

0.00%

-1.00%

-2.00%

-3.00%

-4.00%

-5.00%

-10 -5

0

5

10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85

Temperature, C

Figure 5-6: Typical Signal Deviation vs. Temperature

Vdd = 5.0 Volts, Cx = 10pF, Cs = 5nF - 200nF PPS Film

LQ

17

QT310/R1.03 21.09.03

450

400

350

300

250

200

150

100

50

Sleep Cycles

None

One

Two

Three

Five

0

10

20

30

40

50

60

Sampling Capacitor (nF)

Figure 5-7 Power Consumption vs Cs

at Selected values of Sleep Cycles;

Cx = 10pF, Vdd = 2.0 Volts

900

800

700

600

500

400

300

200

100

0

Sleep Cycles

None

One

Two

Three

Five

Ten

0

10

20

30

40

50

60

Sampling Capacitor (nF)

Figure 5-8 Power Consumption vs Cs

at Selected values of Sleep Cycles;

Cx = 10pF, Vdd = 3.3 Volts

2000

1800

1600

1400

1200

1000

800

Sleep Cycles

None

One

Two

Three

Five

Ten

600

400

200

0

10

20

30

40

50

60

Sampling Capacitor (nF)

Figure 5-9 Power Consumption vs Cs

at Selected values of Sleep Cycles;

Cx = 10pF, Vdd = 5.0 Volts

LQ

18

QT310/R1.03 21.09.03

M

A

F

S1

S

a

A

r

L2

Pin 1

x

m

L1

L

Q

Package type: 8-pin Dual-In-Line

Millimeters

Inches

SYMBOL

Min

6.1

Max

7.11

8.26

10.16

-

Notes

Min

0.24

0.3

Max

0.28

0.325

0.4

Notes

a

A

7.62

9.02

7.62

0.69

0.356

1.14

0.203

2.54

0.38

2.92

-

M

m

Q

L

0.355

0.3

Typical

-

Typical

0.94

0.559

1.78

0.305

-

0.027

0.014

0.045

0.008

0.1

0.037

0.022

0.07

0.012

-

L1

L2

F

BSC

r

-

0.015

0.115

-

S

3.81

5.33

10.9

0.15

0.21

0.43

S1

x

M

a

H

A

φ

e

h

Pin 1

E

F

L

Package type: 8-pin Wide SOIC

Millimeters

Inches

SYMBOL

Min

5.21

7.62

Max

5.41

8.38

Notes

Min

0.205

0.3

Max

0.213

0.33

Notes

a

A

M

F

L

5.16

1.27

5.38

0.203

0.05

0.212

BSC

0.305

0.102

1.78

0.508

0.33

0.012

0.004

0.07

0.02

0.013

0.08

h

H

e

2.03

0.254

0.178

0.508

o

0.007

0.02

o

0.01

0.035

o

E

φ

0.889

o

0

8

0

8

LQ

19

QT310/R1.03 21.09.03

lQ

Patented and patents pending

Corporate Headquarters

1 Mitchell Point

Ensign Way, Hamble SO31 4RF

Great Britain

Tel: +44 (0)23 8056 5600 Fax: +44 (0)23 8045 3939

admin@qprox.com

www.qprox.com

North America

651 Holiday Drive Bldg. 5 / 300

Pittsburgh, PA 15220 USA

Tel: 412-391-7367 Fax: 412-291-1015

The specifications set out in this document are subject to change without notice. All products sold and services supplied by QRG are subject

to our Terms and Conditions of sale and supply of services which are available online at www.qprox.com and are supplied with every order

acknowledgement. QProx, QTouch, QMatrix, QLevel, and QSlide are trademarks of QRG. QRG products are not suitable for medical

(including lifesaving equipment), safety or mission critical applications or other similar purposes. Except as expressly set out in QRG's Terms

and Conditions, no licenses to patents or other intellectual property of QRG (express or implied) are granted by QRG in connection with the

sale of QRG products or provision of QRG services. QRG will not be liable for customer product design and customers are entirely

responsible for their products and applications which incorporate QRG's products.

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